© Semiconductor Components Industries, LLC, 2007
May, 2007 - Rev. 4
1Publication Order Number:
NCP1027/D
NCP1027
High-Voltage Switcher
for Medium Power Offline
SMPS Featuring Low
Standby Power
The NCP1027 offers a new solution targeting output power levels
from a few watts up to 15 W in a universal mains flyback application.
Our proprietary high- voltage technology lets us include a power
MOSFET together with a startup current source, all directly
connected to the bulk capacitor. To prevent lethal runaway in low
input voltage conditions, an adjustable brown- out circuitry blocks
the activity until sufficient input level is reached.
Current- mode operation together with an adjustable ramp
compensation offers superior performance in universal mains
applications. Furthermore, an Over Power Protection pin brings the
ability to precisely compensate all internal delays in high input voltage
conditions and optimize the maximum output current capability.
Protection wise, a timer detects an overload or a short- circuit and
stops all operations, ensuring a safe auto- recovery, low duty cycle burst
operation. An integrated, auto- recovery, Overvoltage Protection
permanently monitors the VCC level and temporarily shuts down the
driving pulses in case of an unexpected feedback loop runaway.
Finally, a great RDS(on) figure makes the circuit an excellent choice
for standby/auxiliary offline power supplies or applications requiring
higher output power levels.
Features
•Built-in 700 V MOSFET with Typical RDS(on) of 5.8 W, TJ = 25°C
•Current-Mode Fixed Frequency Operation: 65 kHz and 100 kHz
•Fixed Peak Current of 800 mA
•Skip-Cycle Operation at Low Peak Currents
•Internal Current Source for Clean and Lossless Startup Sequence
•Auto-Recovery Output Short Circuit Protection with Timer-Based
Detection
•Auto-Recovery Overvoltage Protection with Auxiliary Winding
Operation
•Programmable Brown-Out Input for Low Input Voltage Detection
•Programmable Over Power Protection
•Input to Permanently Latchoff the Part
•Internal Frequency Jittering for Improved EMI Signature
•Extended Duty Cycle Operation to 80% Typical
•No-Load Input Standby Power of 85 mW @ 265 Vac
•500 mW Loaded, Input Power of 715 mW @ 230 Vac
•These are Pb-Free Devices
Typical Applications
•Medium Power AC-DC Adapters for Chargers
•Auxiliary/Standby Power Supplies for ATX and TVS Power Supplies
Reference 230 VAC 90-265 VAC
NCP1027 - 5.8 W25 W* 15 W*
*Typical values, open-frame, 65 kHz version, RqJA < 75°C/W, TA = 50°C.
8-LEAD PDIP
P SUFFIX
CASE 626A
PIN CONNECTIONS
Device Package Shipping*
ORDERING INFORMATION
NCP1027P065G PDIP-8
(Pb-Free)
50 Units / Rail
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MARKING
DIAGRAM
XXX = 65 or 100
A = Assembly Location
WL = Wafer Lot
YY = Year
WW = Work Week
G = Pb-Free Package
(Top View)
Ramp Comp.
VCC
Brown-Out
FB Drain
P1027PXXX
AWL
YYWWG
GND
OPP
*For additional information on our Pb-Free strategy
and soldering details, please download the
ON Semiconductor Soldering and Mounting
Techniques Reference Manual, SOLDERRM/D.
NCP1027P100G PDIP-8
(Pb-Free)
50 Units / Rail
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85-265 VAC
+
BO
+
OVP
+
1
2
3
45
7
8
NCP1027
OPP*
+
Vout
GND
Figure 1. Typical Application
Ramp
Comp.*
*Optional component
PIN FUNCTION DESCRIPTION
Pin No. Symbol Function Description
1 VCC Powers the Internal
Circuitry
This pin is connected to an external capacitor of typically 22 mF. The VCC
includes an active shunt which serves as an auto-recovery overvoltage
protection.
2Ramp Comp. Ramp Compensation
in CCM
To extend the duty cycle operation in Continuous Conduction Mode (CCM),
pin 3 offers the ability to inject ramp compensation in the controller. If unused,
short this pin to VCC.
3 Brown-Out Brown-Out and
Latchoff Input
By monitoring the bulk level via a resistive network, the circuit protects itself
from low mains conditions. If an external event brings this pin above 4.0 V,
the part fully latches off.
4 FB Feedback Signal Input By connecting an optocoupler to this pin, the peak current setpoint is adjusted
accordingly to the output power demand.
5 Drain Drain Connection The internal drain power switch circuit connection.
- - - This unconnected pin ensures adequate creepage distance.
7 OPP Over Power Protection Driving this pin reduces the power supply capability in high line conditions. If
no Over Power Protection is needed, short this pin to ground.
8 GND The IC Ground -
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VCC
Vclamp
Auto-Recovery OVP
+
-
+
-
+
VCC
Mngt
Fault
UVLOs
4 V rst
VDD
IC1
GND
20 ms RC
SQ
Q
R
LEB
Timer
Ip Flag
+
-
VCC < 4 V
Reset
IBO
+
VBO
65 kHz or
100 kHz
CLOCK
SQ
Q
R
Jittering
Vdd
BO
Ramp
Comp.
+
25%
of lp
+
-
Vdd
RFB
FB
Skip
+
-
Soft-Start
Ip Flag
+
-
OPP
Vcc
Drain
UVLO
Figure 2. Internal Block Diagram
+
Vlatch
Icomp
Max Ip Selection
Over Power
Protection Input
Ramp Compensation
50 ms RC
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MAXIMUM RATINGS
Rating Symbol Value Unit
Power Supply Voltage on all Pins, Except Pin 5 (Drain) VCC -0.3 to 10 V
Drain Voltage BVdss -0.3 to 700 V
Drain Current Peak During Transformer Saturation IDS(pk) 1.8 A
Maximum Current into Pin 1 when Activating the 8.7 V Active Clamp I_VCC 15 mA
Thermal Resistance, Junction-to-Air – PDIP7 RqJA 100 °C/W
Thermal Resistance, Junction-to-Air – PDIP7 with 1.0 cm@ of 35 m Copper Area RqJA 75 °C/W
Maximum Junction Temperature TJMAX 150 °C
Storage Temperature Range --60 to +150 °C
ESD Capability, Human Body Model (HBM) (All Pins Except HV) - 2.0 kV
ESD Capability, Machine Model (MM) - 200 V
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
1. This device series contains ESD protection and exceeds the following tests:
Human Body Model 2000 V per Mil-Std-883, Method 3015.
Machine Model Method 200 V.
2. This device contains latchup protection and exceeds 100 mA per JEDEC Standard JESD78.
ELECTRICAL CHARACTERISTICS (For typical values TJ = 25°C, for min/max values TJ = 0°C to +125°C, Max TJ = 150°C,
VCC = 8.0 V, unless otherwise noted.)
Characteristic Pin Symbol Min Typ Max Unit
SUPPLY SECTION AND VCC MANAGEMENT
VCC Increasing Level at which the Switcher Starts to Operate 1 VCCON 7.9 8.5 8.9 V
VCC Decreasing Level at which the Switcher Stops Operation 1 VCC(min) 6.7 7.2 7.9 V
Hysteresis between VCCON and VCC(min) - VCChyste - 1.2 - V
Offset Voltage above VCCON at which the Internal Clamp Activates 1 VCCclamp 140 200 300 mV
VCC Voltage at which the Internal Latch is Reset 1 VCCreset - 4.0 - V
Internal IC Consumption, MOSFET Switching at 65 kHz or 100 kHz 1 ICC1 - 1.4 1.9 mA
POWER SWITCH CIRCUIT
Power Switch Circuit On-State Resistance
NCP1027 (Id = 100 mA)
TJ = 25°C
TJ = 125°C
5 RDS(on)
-
-5.8
9.8
7.0
11
W
Power Switch Circuit and Startup Breakdown Voltage
(ID(off) = 120 mA, TJ = 25°C)
5 BVdss 700 - - V
Power Switch and Startup Breakdown Voltage Off-State
Leakage Current
TJ = 25°C (Vds = 700 V)
TJ = 125°C (Vds = 700 V) 5
5
Idss(OFF)
-
-50
30
-
-
mA
Switching Characteristics (RL = 50 W, Vds Set for
Idrain = 0.7 x Ilim)
Turn-on Time (90%-10%)
Turn-of f Time (10%-90%) 5
5
ton
toff
-
-35
35
-
-
ns
ns
INTERNAL STARTUP CURRENT SOURCE
High-Voltage Current Source, VCC = VCCON – 200 mV 1 IC1 3.5 6.0 8.0 mA
High-Voltage Current Source, VCC = 0 1 IC2 350 650 900 mA
VCC Transition Level for IC1 to IC2 Toggling Point 1 VCCTh - 1.3 - V
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ELECTRICAL CHARACTERISTICS (continued) (For typical values TJ = 25°C, for min/max values TJ = 0°C to +125°C,
Max TJ = 150°C, VCC = 8.0 V, unless otherwise noted.)
Characteristic Pin Symbol Min Typ Max Unit
CURRENT COMPARATOR
Maximum Internal Current Setpoint, Pin 4 Open, TJ = 25°C,
FSW = 65 kHz (Note 3)
- Ipeak_27_CS_
65 k
720 800 880 mA
Final Switch Current with a Primary Slope of 200 mA/ms,
FSW = 65 kHz (Note 4)
- Ipeak_27_SW_
65 k
- 820 - mA
Maximum Internal Current Setpoint, Pin 4 Open, TJ = 25°C,
FSW = 100 kHz (Note 3)
- Ipeak_27_CS_
100 k
720 800 880 mA
Final Switch Current with a Primary Slope of 200 mA/ms,
FSW = 100 kHz (Note 4)
- Ipeak_27_SW_
100 k
- 820 - mA
Setpoint Decrease for a Pin 7 Injected Current of 40 mA, TJ = 25°C 7 IOPP - 23 - %
Voltage Level in Pin 7 at which OPP Starts to Operate 7 IOPPtripV - 1.5 - V
Soft-Start Duration - TSS - 1.0 - ms
Propagation Delay from Current Detection to Drain OFF State - Tprop - 100 - ns
Leading Edge Blanking Duration - TLEB - 200 - ns
INTERNAL OSCILLATOR
Oscillation Frequency (Note 5)
65 kHz Version, TJ = 25°C
- fOSC 58.5 65 71.5
kHz
Oscillation Frequency (Note 5)
100 kHz Version, TJ = 25°C
- fOSC 90 100 110
kHz
Frequency Jittering in Percentage of fOSC - fJitter -"6.0 - %
Jittering Swing Frequency - fswing - 300 - Hz
Maximum Duty Cycle - Dmax 74 80 87 %
FEEDBACK SECTION
Internal Pullup Resistor 4 Rupp - 16 - kW
Ramp Compensation Level on Pin 1 – Rramp = 100 kW2 Rlevel - 2.75 - V
SKIP CYCLE GENERATION
Internal Skip Mode Level, in Percentage of Maximum Peak Current - Iskip - 25 - %
PROTECTIONS
Brown-Out Level 3 VBO 510 570 620 mV
Brown-Out Hysteresis Current, TJ = 25°C (Note 3) 3 IBOhyste 10 11.5 13 mA
Brown-Out Hysteresis Current, TJ = 0°C to 125°C 3 IBOhyste - 10 - mA
Fault Validation further to Error Flag Assertion - TimerON 40 55 - ms
OFF Phase in Fault Mode - TimerOFF - 440 - ms
Latching Voltage on Brown-Out Pin 3 Vlatch 3.15 3.5 3.85 V
Latch Input Integrating Filter Time Constant 3 TdelBOL - 20 - ms
OVP Integrating Filter Time Constant - TdelOVP - 50 - ms
VCC Current at which the Switcher Stops Pulsing 1 IOVP 6.0 8.5 11 mA
TEMPERATURE MANAGEMENT
Temperature Shutdown - TSD 160 - - °C
Hysteresis in Shutdown - - - 40 - °C
3. See characterization curves for full temperature span evolution.
4. The final switch current is: Ipeak_2X_CS + Tprop x Vin / Lp, with Vin the input voltage and Lp the primary inductor in a flyback.
5. Oscillator frequency is measured with disabled jittering.
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11
10
9
8
7
6140120100806040200-20
TEMPERATURE (°C)
IOVP (mA)
7.9
8.0
8.1
8.2
8.3
8.4
8.5
8.6
8.7
8.8
8.9
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
VCCON (V)
TEMPERATURE (°C)
6.7
6.8
6.9
7.0
7.1
7.2
7.3
7.4
7.5
7.6
7.7
7.8
7.9
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
VCCMIN
(
V
)
140120100806040200-20
TEMPERATURE (°C)
VCCClamp (V)
0.14
0.16
0.18
0.20
0.22
0.24
1.0
1.2
1.4
1.6
1.8
-20 0 20 40 60 80 100 120 140
ICC1 (mA)
350
400
450
500
550
600
650
700
750
800
850
900
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
IC2 (mA)
Figure 3. Figure 4.
Figure 5. Figure 6.
Figure 7. Figure 8.
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3.5
4.0
4.5
5.0
5.5
6.0
6.5
7.0
7.5
8.0
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
IC1 (mA)
59.0
60.0
61.0
62.0
63.0
64.0
65.0
66.0
67.0
68.0
69.0
70.0
71.0
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
Fosc (kHz)
75.0
77.0
79.0
81.0
83.0
85.0
87.0
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
Dmax (%)
Figure 9. Figure 10.
Figure 11. Figure 12.
0
20
40
60
80
100
120
TEMPERATURE (°C)
Fosc (kHz)
TEMPERATURE (°C)
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
510
520
530
540
550
560
570
580
590
600
610
VBO (mV)
9.0
9.5
10.0
10.5
11.0
11.5
12.0
12.5
13.0
-20 0 20 40 60 80 100 120 140
IBO HYSTERESIS (mA)
Figure 13. Figure 14.
-40 -20 0 20 40 60 80 100 120 140
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-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
Vlatch (V)
3.2
3.3
3.4
3.5
3.6
3.7
3.8
Figure 15.
-20 0 20 40 60 80 100 120 14
0
TEMPERATURE (°C)
RDS(on) @ ID = 100 mA (W)
2
3
4
5
6
7
8
9
10
11
Figure 16.
720
740
760
780
800
820
840
860
880
TEMPERATURE (°C)
-20 0 20 40 60 80 100 120 140
I
pea
k
(
m
A)
Figure 17.
16
18
20
22
24
26
28
30
-20 0 20 40 60 80 100 120 14
0
TEMPERATURE (°C)
Iopp (%)
Ipin 7 = 40 mA
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
Tleb + Tpropdelay (ns)
200
220
240
260
280
300
320
340
360
380
400
Figure 18.
Figure 19.
2.5
2.6
2.7
2.8
TEMPERATURE (°C)
-20 0 20 40 60 80 100 120 14
0
RAMP COMPENSATION LEVEL (V)
Figure 20.
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-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
TimerON (ms)
50
52
54
56
58
60
62
64
66
68
70
Figure 21.
-20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
Idss OFF (mA)
10
20
30
40
50
60
70
80
90
100
Figure 22.
0%
10%
20%
30%
40%
50%
60%
70%
80%
90%
100%
250200150100500
140°C
125°C
85°C
25°C0°C
-20 °C
-40 °C
IOPP (mA)
IPEAK REDUCTION (%)
Figure 23. Ipeak Reduction = F(lopp, @ temperature)
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APPLICATION INFORMATION
Introduction
The NCP1027 offers a complete current-mode control
solution and enhances the NCP101X series. The
component integrates everything needed to build a rugged
and low-cost Switch-Mode Power Supply (SMPS)
featuring low standby power.
•Current-Mode Operation: The controller uses a
current-mode control architecture, which, together
with an adjustable ramp compensation circuitry,
ensures efficient and stable continuous or
discontinuous conduction designs.
•700 V–5.8 W Power Switch Circuit: Due to
ON Semiconductor Very High Voltage Integrated
Circuit technology, the circuit hosts a high-voltage
power switch circuit featuring a 5.8 W RDS(on) – TJ =
25°C. This value lets the designer build a 15 W
power supply operated on universal mains as
long as sufficient copper area exists to lower the
junction-to-ambient thermal resistance. An internal
current source delivers the startup current, necessary
to crank the power supply.
•Short-Cir cuit Protection: By permanently
monitoring the feedback line activity, the circuit is
able to detect the presence of a short-circuit,
immediately reducing the output power for a total
system protection. A 55 ms timer is started as soon as
the feedback pin asks for the maximum peak current.
At the end of this timer, if the fault is still present,
then the device enters a safe, auto-recovery burst
mode, affected by a fixed 440 ms recurrence. Once the
short has disappeared, the controller resumes and goes
back to normal operation. The timer duration is fully
independent from the VCC capacitor value.
•Over Power Protection: A possibility exists to reduce
the maximum output power capability in high line
conditions. A simple two resistor network wired to the
bulk capacitor will program the maximum current
reduction for a given input voltage (down to 20% of
the maximum peak current).
•Fail-Safe Optocoupler/Overvoltage Protection: As
the auxiliary winding is connected to the VCC pin, an
internal active clamp connected between VCC and
ground limits the supply dynamics to 8.7 V. In case
the current injected in this clamp exceeds a level of
6.0 mA (minimum), the controller immediately stops
switching and waits a full timer period before
attempting to restart. If the default is gone, the
controller resumes operation. If the default is still
there, e.g. a broken optocoupler, the controller protects
the load through a safe burst mode.
•Brown-Out Input: A fraction of the input voltage
appears on pin 3, due to a resistive divider. If the
mains drops below a level adjusted by this resistive
divider, the circuit does not switch. As soon as the
mains goes back within its normal range, the device
resumes operation and operates normally. By adjusting
the bridge resistors, it becomes possible to set the
brown-out levels (on and off) independently.
•Latchoff: Pin 3 also welcomes a comparator who
offers a way to fully latch the controller. If an external
event (e.g. an overtemperature) brings the brown-out
pin above 3.5 V, the circuit stays permanently off
until the user cycles its VCC down, for instance by
unplugging the converter from the mains outlet.
•Frequency Jittering: The internal clock receives a
low frequency modulation which helps smoothing the
power supply EMI signature.
•Soft-Start: A 1.0 ms soft-start ensures a smooth
startup sequence, reducing output overshoots.
•Skip Cycle: If SMPS naturally exhibit a good
efficiency at nominal load, they begin to be less
efficient when the output power demand diminishes.
By skipping unneeded switching cycles, the NCP1027
drastically reduces the power wasted during light load
conditions. Experiments carried over the 5.0 V/2.0 A
demonstration board reveal a standby power at
no-load and 265 Vac of 85 mW and an efficiency for
500 mW output power of 64% at 230 Vac.
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Startup Sequence
The NCP1027 includes a high-voltage startup circuitry,
directly deriving current from the bulk line to charge the
VCC capacitor. Figure 24 details the simplified internal
arrangement.
Vbulk
I1
RVCC
1
I2
+CVCC
Iclamp
Vz = 8.7 V
Iclamp > 6 mA
→ OVP fault
ICC1
+
-
VCCon
VCCoff
+
5
8
+
←
IC1
Figure 24. Internal Arrangement of the Startup Circuitry
When the power supply is first connected to the mains
outlet, the internal current source is biased and charges up
the VCC capacitor. When the voltage on this VCC capacitor
reaches the VCCON level (typically 8.5 V), the current
source turns off, reducing the amount of power being
dissipated. At this time, the VCC capacitor only supplies the
controller, and the auxiliary supply should take over before
VCC collapses below VCC(min). This VCC capacitor, CVCC,
must therefore be calculated to hold enough energy so that
VCC stays above VCC(min) (7.3 V typical) until the
auxiliary voltage fully takes over.
An auxiliary winding is needed to maintain the VCC in
order to self-supply the switcher. The VCC capacitor has
only a supply role and its value does not impact other
parameters such as fault duration or the frequency sweep
period for instance. As one can see in Figure 24, an internal
active Zener diode, protects the switcher against lethal VCC
runaways. This situation can occur if the feedback loop
optocoupler fails, for instance, and you would like to
protect the converter against an over voltage event. In that
case, the internal current increase incurred by the VCC
rapid growth triggers the over voltage protection (OVP)
circuit and immediately stops the output pulses for 440 ms.
Then a new startup attempt takes place to check whether
the fault has disappeared or not. The OVP paragraph gives
more design details on this particular section.
The VCC capacitor can be calculated knowing a) the
amount of energy that needs to be stored; b) the time it
takes for the auxiliary voltage to appear, and; c) the current
consumed by the controller at that time. For a better
understanding, Figure 25 shows how the voltage evolves
on the VCC capacitor upon startup.
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Figure 25. A typical startup sequence showing the VCC capacitor
voltage evolution versus time.
Suppose our power supply takes 10 ms (tstartup) to bring
the output voltage to its target value. We know that the
switcher consumption is around 2.0 mA (ICC1). Therefore,
we can calculate the amount of capacitance we need, to
hold VCC above 7.5 V at least for 10 ms while delivering
2.0 mA:
CwICC1tstartup
DVCC or, by replacing with the above values,
Cw2m·10m
1w20mF then select a 33 mF for the VCC
capacitor.
Fault Condition – Short-Circuit on VCC
In some fault situations, a short-circuit can purposely
occur between VCC and GND. In high line conditions
(VHV = 370 VDC) the current delivered by the startup
device will seriously increase the junction temperature. For
instance, since IC1 equals 3.0 mA (the min corresponds to
the highest TJ), the device would dissipate 370 3 m =
1.1 W. To avoid this situation, the controller includes a
novel circuitry made of two startup levels, IC1 and IC2. At
powerup, as long as VCC is below a 1.3 V level, the source
delivers IC1 (around 650 mA typical), then, when VCC
reaches 1.3 V, the source smoothly transitions to IC2 and
delivers its nominal value. As a result, in case of
short-circuit between VCC and GND, the power dissipation
will drop to 370 650 m = 240 mW. Figure 26 portrays
this particular behavior.
Figure 26. The startup source now features a
dual-level startup current.
The first startup period is calculated by the formula
C V = I t, which implies a 33 m 1.3/650 m = 66 ms
startup time for the first sequence (t1). The second
sequence (t2) is obtained by toggling the source to 4.0 mA
with a delta V of VCCON – VCCth = 8.5 – 1.5 = 7.0 V,
which finally leads to a second startup time of
7 33 m/6.0 m = 39 ms. The total startup time becomes
66 m + 39 m = 105 ms as a typical value. Please note that
this calculation is approximated by the presence of the knee
in the vicinity of the transition.
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Fault Condition – Output Short-Circuit
As soon as VCC reaches VCCON, drive pulses are
internally enabled. If everything is correct, the auxiliary
winding increases the voltage on the VCC pin as the output
voltage rises. During the start-sequence, the controller
smoothly ramps up the peak current to Imax setting, e.g.
Ipeak_HI, which is reached after a typical period of 1.0 ms.
As soon as the peak current setpoint reaches its maximum
(during the startup period but also anytime an overload
occurs), an internal error flag is asserted, Ipflag, indicating
that the system has reached its maximum current limit set
point (Ip = Ip max). The assertion of this flag triggers a
55 ms counter. If at counter completion Ipflag remains
asserted, all driving pulses are stopped and the part stays off
during eight periods of 55 ms (440 ms). A new attempt to
restart occurs and will last 55 ms providing the fault is still
present. If the fault still affects the output, a safe burst mode
is entered, affected by a low duty-cycle operation (11%).
When the fault disappears, the power supply quickly
resumes operation. Figure 27 depicts this particular mode.
Figure 27. In case of short-circuit or overload, the NCP1027 protects itself and the power supply via a low
frequency burst mode. The VCC is maintained by the current source and self-supplies the controller.
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In Figure 27, one can see that the VCC is still alive,
testifying for a badly coupled power secondary and
primary auxiliary windings. Some situations exist where an
output short-circuit make the auxiliary winding collapse
before the timer completion. In this particular case, the
Undervoltage Lock Out (UVLO) circuitry has the priority
and safely cuts off all driving pulses. Figure 28 describes
this variation.
Figure 28. The auxiliary winding collapses in presence of a short-circuit. Pulses are immediately stopped
as VCC crosses the minimum operating voltage, VCC(min).
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Fault Condition – Output Too Low
This particular mode of operation occurs when the
feedback is ensured by a two-loop control imposing either
constant output voltage (CV) or constant output current
(CC), for instance in a battery charger. In CC mode, the
output voltage falls down below the original target but the
feedback loop is kept closed by the CC controller. For that
reason, the controller becomes un-able to detect a real
output short-circuit since Ipflag will never be asserted.
Due to a good winding coupling, the primary side auxiliary
collapsing will ensure a proper fault detection via the
UVLO internal circuit. Figure 29 depicts this operating
way.
Figure 29. In this particular case, the output goes low but the timer is not started since the FB pin is still held
by the optocoupler. Due to the UVLO circuit, the controller safely stops operation at VCC = VCC(min).
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Fault Condition – Low Input Voltage
The NCP1027 includes a brown-out circuitry able to
protect the power supply in case of low input voltage
conditions. Figure 30 shows how internally the NCP1027
monitors the voltage image of the bulk capacitor. Below a
given level, the controller blocks the driving pulses, above
it, it authorizes them. The internal circuitry, depicted by
Figure 30a, offers a way to observe the high-voltage (HV)
rail. A resistive divider made of Rupper and Rlower, brings
a portion of the HV rail on pin 3. Below the turn-on level,
the 10 mA current source IBO is off. Therefore, the turn-on
level solely depends on the division ratio brought by the
resistive divider.
←
+
-BO
ON/OFF
VDD
IBO
BO
Vbulk
Rupper
Rlower +
VBO
1vin 2vcmp
20.0u 60.0u 100u 140u 180u
Time in Seconds
0
40.0
80.0
120
160
0
4.00
8.00
12.0
16.0
Plot1
2
1
Vcmp Volts
Vin in Volts
Vbulk = 100 V Vbulk = 70 V
Figure 30a. The internal brown-out
configuration with an offset current source.
Figure 30b. Simulation results for 100/70 ON/OFF levels.
Figure 30.
To the contrary, when the internal BO signal is high, the IBO source is activated and creates an hysteresis. As a result,
it becomes possible to select the turn-on and turn-off levels via a few lines of algebra.
IBOisOff
(eq. 1)
V())+Vbulk1 Rlower
Rlower )Rupper
IBOisOn
(eq. 2)
V())+Vbulk2 Rlower
Rlower )Rupper )IBO ǒRlower Rupper
Rlower )RupperǓ
We can now extract Rlower from Equation 1 and plug it into Equation 2, then solve for Rupper :
Rupper +Rlower Vbulk1- VBO
VBO
Rlower +VBO Vbulk1-V
bulk2
IBO (Vbulk1- VBO)
If we decide to turn-on our converter for Vbulk1 equals 100 V and turn it off for Vbulk2 equals 70 V, then we obtain:
Rupper = 3.0 MW
Rlower = 18 kW
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The bridge power dissipation is 3302/3.018 Meg =
36 mW in nominal high-line operation. Figure 30b
simulation result confirms our calculations.
Figure 31 describes signal variations during a brown-out
condition. Please note that output pulses only reappear
when VCC reaches VCC(ON), ensuring a clean startup
sequence. As in fault mode conditions, the startup source
is activated on and off and self-supplies the controller in a
Dynamic Self-Supply (DSS) mode.
Figure 31. Signal Evolution During a Brown-Out Condition
Depending on input surge tests, it might be necessary to
wire a filtering capacitor between BO and GND (close to
the circuit) to avoid adversely triggering the internal latch
(unless this is a wanted feature) when the pulse train
appears.
Latchoff Protection
There are some situations where the converter shall be
fully turned-off and stay latched. This can happen in the
presence of a secondary overvoltage (the feedback loop is
drifting) or when an overtemperature is detected.
Secondary monitoring is usually implemented when the
coupling between auxiliary and power windings does not
lead to a precise primary detection, hence the
auto-recovery OVP on pin 1 would not satisfy the precision
requirements. Due to the addition of a comparator on the
BO pin, a simple external circuit can lift up this pin above
VLATCH and permanently disable pulses. The VCC needs
to be cycled down below 3.5 V typically to reset the
controller.
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+
-
Vout +
Vlatch
NTC
VCC
Q1
Vbulk
Rlower
Rupper
BO
20 ms
RC To permanent
latch
←
IBO
VDD
+
-BO
+
VBO
Figure 32. Adding a comparator on the BO pin offers a way to latch-off the controller.
In Figure 32, Q1 is blocked and does not bother the BO
measurement as long as the NTC and the optocoupler are
not activated. As soon as the secondary optocoupler senses
an OVP condition, or the NTC reacts to a high ambient
temperature, Q1 base is brought to ground and the BO pin
goes up, permanently latching off the controller. Figure 33
depicts the converter behavior in case of total latch-off.
Figure 33. If the BO pin is lifted up to VLATCH, the controller permanently latches off.
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Auto-Recovery Overvoltage Protection
The particular NCP1027 arrangement offers a simple
way to prevent output voltage runaway when the
optocoupler fails. As Figure 34 shows, an active Zener
diode monitors and protects the VCC pin. Below its
equivalent breakdown voltage, that is to say 8.7 V typical,
no current flows in it. If the auxiliary VCC pushes too much
current inside the Zener, then the controller considers an
OVP situation and stops the pulses. Figure 34 shows that
the insertion of a resistor (Rlimit) between the auxiliary DC
level and the VCC pin is mandatory a) not to damage the
internal 8.7 V Zener diode during an overshoot for instance
(absolute maximum current is 15 mA) b) to implement the
fail-safe optocoupler protection (OVP) as offered by the
active clamp. Please note that there cannot be bad
interaction between the clamping voltage of the internal
Zener and VCCON since this clamping voltage is actually
built on top of VCCON with a fixed amount of offset
(200 mV typical). Rlimit should be carefully selected to
avoid triggering the OVP as we discussed, but also to avoid
disturbing the VCC in low/light load conditions. The
following details how to evaluate the Rlimit value.
Self-supplying controllers in extremely low standby
applications often puzzles the designer. Actually, if an
SMPS operated at nominal load can deliver an auxiliary
voltage of an arbitrary 16 V (Vnom), this voltage can drop
below 10 V (Vstby) when entering standby. This is because
the recurrence of the switching pulses expands so much,
that the low frequency refueling rate of the VCC capacitor
is not enough to keep a proper auxiliary voltage. Figure 35
portrays a typical scope shot of an SMPS entering deep
standby (output unloaded). Thus, care must be taken when
calculating Rlimit 1) to not trigger the VCC overcurrent latch
(by injecting 6.0 mA into the active clamp – always use the
minimum value for worse case design) in normal operation
but 2) not to drop too much voltage over Rlimit when
entering standby. Otherwise, the converter will enter burst
mode as it will sense an UVLO condition. Based on these
recommendations, we are able to bound Rlimit between two
equations:
(eq. 3)
Vnom-V
clamp
Itrip vRlimit vVstby-V
CCON
ICC1
Where:
Vnom is the auxiliary voltage at nominal load.
Vstdby is the auxiliary voltage when standby is entered.
Itrip is the current corresponding to the nominal operation.
It thus must be selected to avoid false tripping in overshoot
conditions. Always use the minimum of the specification
for a robust design.
ICC1 is the controller consumption. This number slightly
decreases compared to ICC1 from the spec since the part
in standby does almost not switch. It is around 1.0 mA for
the 65 kHz version.
VCC(min) is the level above which the auxiliary voltage
must be maintained to keep the controller away from the
UVLO trip point. It is good to obtain around 8.0 V in order
to offer an adequate design margin, e.g. to not reactivate the
startup source (which is not a problem in itself if low
standby power does not matter).
←
+
-
+
-
+
VCCON = 8.5 V
VCC(min) = 7.5 V
Startup
Source
Drain
+
+
-
Vclamp = 8.7 V Typ.
Latch
+I > 6 mA
Ground
VCC
+CVCC
+CAUX Laux
Rlimit D1
Figure 34. A more detailed view of the NCP1027 offers better
insight on how to properly wire an auxiliary winding.
Since Rlimit shall not bother the controller in standby, e.g.
keep Vauxiliary to around 8.0 V (as selected above), we
purposely select a Vnom well above this value. As
explained before, experience shows that a 40% decrease
can be seen on auxiliary windings from nominal operation
down to standby mode. Let's select a nominal auxiliary
winding of 20 V to offer sufficient margin regarding 8.0 V
when in standby (Rlimit also drops voltage in standby…).
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Plugging the values in Equation 3 gives the limits within
which Rlimit shall be selected:
20- 8.7
6m vRlimit v12- 8
1m , thatissay : 1.8kWtRlimit
t4kW.
to
If we design a 65 kHz power supply delivering 12 V, then
the ratio between auxiliary and power must be: 12/20 = 0.6.
The OVP latch will activate when the clamp current
exceeds 6.0 mA. This will occur when Vauxiliary grows up
to:
1. 8.7 + 1.8 k (6 m + 1.8 m) ≈ 23 V for the first
boundary (Rlimit = 1.8 kW).
2. 8.7 + 4 k (6 m + 1.8 m) ≈ 40 V for the second
boundary (Rlimit = 4.0 kW).
Due to a 0.6 ratio between the auxiliary VCC and the
power winding, the auxiliary OVP will be seen as a lower
overshoot on the real output:
1. 23 0.6 ≈ 13.8 V
2. 40 0.6 ≈ 24 V
As one can see, tweaking the Rlimit value will allow the
selection of a given overvoltage output level. Theoretically
predicting the auxiliary drop from nominal to standby is an
almost impossible exercise since many parameters are
involved, including the converter time constants. Fine
tuning of Rlimit thus requires a few iterations and
experiments on a breadboard to check the auxiliary voltage
variations but also the output voltage excursion in fault.
Once properly adjusted, the fail-safe protection will
preclude any lethal voltage runaways in case a problem
would occur in the feedback loop.
Figure 35. The burst frequency becomes so low
that it is difficult to keep an adequate level on the
auxiliary VCC.
> 30 ms
Figure 36 describes the main signal variations when the
part operates in auto-recovery OVP.
Figure 36. If the VCC current exceeds a certain threshold, an auto-recovery protection is
activated and protects the design.
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Improving the Precision in Auto-Recovery OVP
Given the OVP variations the internal trip current
dispersion incur, it is sometimes more interesting to
explore a different solution, improving the situation to the
cost of a minimal amount of surrounding elements.
Figure 37 shows that adding a simple Zener diode on top
of the limiting resistor, offers a better precision since what
matters now is the internal 8.7 V VCC breakdown, plus the
Zener voltage. A resistor in series with the Zener diodes
keeps the maximum current in the VCC pin below the
maximum rating of 15 mA just before tripping the OVP.
Ground
VCC
+Laux
Rlimit
D1
Figure 37. A simple Zener diode added in
parallel with Rlimit , allows for a better
precision OVP.
Figure 38. Internal logic blocks take a certain amount
of time before shutting off the driving pulses in
presence of an overcurrent event.
Over Power Compensation
Over Power Compensation or Protection (OPP)
represents a way to limit the effects of the propagation
delay when the converter is supplied from its highest input
voltage. The propagation delay naturally extends the
power capability of any current-limited converter.
Figure 38 explains why. The main parameter is the on
slope, that is to say, the pace at which the inductor current
grows-up when the power switch closes. For a flyback
controller, the slope is given by:
Son +Vin
Lp(eq. 4)
where Lp is the transformer magnetizing/primary
inductance and V
in, the input voltage.
As the internal logic takes some time to react, the switch
gate shutdown does not immediately occur when the
maximum power limit is detected (just before activating
the overload protection circuit). Clearly speaking, it can
take up to 100 ns for the NCP1027 current sense
comparator to propagate through the various logical gates
before reaching the power switch and finally shutting it off.
This is the well-known propagation delay noted tprop.
Unfortunately, during this time, the current keeps growing
as Figure 38 depicts. The peak current will therefore be
troubled by this propagation delay. The formula to obtain
the final value is simply:
Ipeak, final +Vin
Lpt (eq. 5)
)Ipeak, max
prop
At low line, Son is relatively low and does not bother the
final peak value. The situation differs at high line and
induces a higher peak current. Therefore, the power supply
output power capability increases with the input voltage.
Let us a take a look at a simple example. Suppose the peak
current is 700 mA:
Lp = 1.0 mH
Vin lowline = 100 Vdc
Vin highline = 350 Vdc
Ipeak,max = 700 mA
tprop = 100 ns
Pout +1
2I2
peak, final FSWLph(eq. 6)
Where: Fsw is the switching frequency and h the efficiency.
Usually h is bigger in high line conditions than in low line
conditions. This formula is valid for a Discontinuous
Conduction Mode flyback.
From Equation 5, we can calculate the final peak current
in both conditions:
Ipeak,final = (100/1m) x 100n + 700m = 710 mA at low line.
Ipeak,final = (350/1m) x 100n + 700m = 735 mA at high line.
From Equation 6, we can have an idea of the maximum
output power capability again, in both conditions with
respective low and high line efficiency numbers of 78%
and 82% for instance:
Pout,lowline = 0.5 0.712 1m 65k 0.78 = 12.8 W
Pout,highline = 0.5 0.7352 1m 65k 0.82 = 14.4 W
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This difference might not be seen as a problem, but some
design specifications impose stringent conditions on the
maximum output current capability, regardless the line
input. Hence the need for an OPP input…
Since we want to limit the power to 12.8 W at high line,
let us calculate the needed peak current:
From equation 6: Ipeak +2Pout
FSWLph
Ǹ = 693 mA to
deliver 12.8 W at high line.
Compared to our 735 mA, we need to decrease the
setpoint by 6% roughly when V
in equals 350 Vdc.
The NCP1027 hosts a special circuitry looking at the
couple voltage/current present on pin 7. Figure 39 shows
how to arrange components around the controller to obtain
Over Power Protection.
Current
Setpoint
Over Power
Protection
OPP
Bulk
ROPPU
ROPPL
GND
Figure 39. A resistive network reduces the
power capability in high-line conditions.
First, you need to know the required injected current and
the voltage across pin 7 to start activating OPP.
Experiments consist in wiring Figure 39 circuit and
running the power supply in conditions where it must shut
down (e.g. highest input voltage and maximum output
current per specification). For this, ROPPL can be put to
10 kW and ROPPU made of a series string of 4 1.0 MW
resistors plus a 10-turn 1.0 MW potentiometer set at its
maximum value. An amp-meter is inserted in series with
pin 7 and a volt-meter monitors its voltage with respect to
ground. Once the power supply is powered, slowly rotate
the potentiometer and observe both voltage and current
going up at pin 7. At a certain time, as voltage and current
increase, the controller will shut down the power supply.
The current at this time is the one we are looking for.
Suppose these experiments lead to 80 mA with a pin 7
activation voltage of 2.45 V. Final resistor equations are:
VbulkH = 375 Vdc ; the maximum voltage at which OPP
must shut down the controller
VbulkL = 200 Vdc ; the minimum voltage below which
OPP is not activated
IOPP = 80 mA ; the current in pin 7
Vf = 2.45 V ; the voltage of pin 7 at the above
condition
ROPPL +VbulkH-V
bulkL
IOPP(VbulkL-V
f)V
f+27kW(eq. 7)
ROPPH +ROPPL VbulkL-V
f
Vf+2.2MW(eq. 8)
If the OPP feature is not needed for some designs, it is
possible to ground it via a copper wire to the adjacent
ground pin. This can help to develop a larger copper area
in an application where the thermal resistance is an
important parameter.
Ramp Compensation
When operating in Continuous Conduction Mode
(CCM), current-mode power supplies can exhibit
so-called sub-harmonic oscillations. To cure this problem,
the designer must inject ramp compensation. The ramp can
either be added to the current sense information or directly
subtracted from the feedback signal. Figure 40 details the
internal arrangement of the ramp compensation circuitry.
Gate Reset
Ramp
RR
Vp
VDD
IRR
Control
Figure 40. The Internal Feedback Chain and the Ramp Compensation Network
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The principle consists in selecting the RR resistor,
connected from pin 2 to ground, to impose a current IRR in
the transistor collector.
Figure 41. Maximum Peak Current Setpoint
Variations versus Ramp Compensation
The equation to get the right compensation level is the
following:
RR +Vp2.75k
Sa@TSW
(eq. 9)
where Vp, the total voltage swing, equals 2.75 V.
Application example:
Suppose we have the following flyback specifications:
Vout = 5.0 V output voltage
Vf = 1.0 V secondary diode forward drop
@ Iout nominal
Np:Ns = 1:N = 1:0.052 transformer turn ratio
Lp = 3.8 mH primary inductance
We can calculate the off slope, the one actually needed
to evaluate Sa, by reflecting the output voltage over the
primary inductance. The slope is projected over a complete
switching period. Here, we use a 65 kHz part.
Soff +Vout )Vf
NLpT
SW +6 15u
0.052 3.8m +455mAń15ms
(eq. 10)
Due to the internal sense arrangement, this current slope
will become a voltage slope having a value of:
SȀoff +455m 0.375 +170mVń15ms(eq. 11)
If we chose 50% of this downslope, then the final
compensation ramp will present a slope of:
Sa+170m
2+85mVń15ms(eq. 12)
We then have:
RR +Vp2.75k
Sa@TSW +2.75 2.75k
85m +89kW(eq. 13)
In the above calculations, the internal ESD resistor has
purposely been omitted to avoid bringing in another
variable. In case no ramp compensation is required, pin 2
must be tied to VCC, the adjacent pin.
Soft-Start
The NCP1027 features a 1.0 ms soft-start, which
reduces the power-on stress, but also contributes to lower
the output overshoot. Figure 42 shows a typical operating
waveform. The NCP1027 features a novel patented
structure which offers a better soft-start ramp, almost
ignoring the startup pedestal inherent to traditional
current-mode supplies.
Figure 42. 1.0 ms Soft-Start Sequence
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Jittering
Frequency jittering is a method used to soften the EMI
signature by spreading the energy in the vicinity of the main
switching component. The NCP1027 offers a "6%
deviation of the nominal switching frequency. The sweep
sawtooth is internally generated and modulates the clock
up and down with a fixed frequency of 300 Hz. Figure 43
shows the relationship between the jitter ramp and the
frequency deviation. It is not possible to externally disable
the jitter.
65kHz
68.9kHz
61.1kHz
Jitter ramp
Internal
sawtooth
adjustable
Figure 43. Modulation Effects on the Clock Signal by the
Jittering Sawtooth
Skip-Cycle
Skip cycle offers an efficient way to reduce the standby
power by skipping unwanted cycles at light loads.
However, the recurrent frequency in skip often enters the
audible range and a high peak current obviously generates
acoustic noise in the transformer. The noise takes its origins
in the resonance of the transformer mechanical structure
which is excited by the skipping pulses. A possible
solution, successfully implemented in the NCP1200 series,
also authorizes skip cycle but only when the power demand
as dropped below a given level. This is what Figure 44
shows, as implemented on the NCP1027.
0
Skip cycle
current limit
Nominal peak
current
Figure 44. Low Peak Current Skip Cycle Guarantees Noise-Free Operation
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5.0 V/3.0 A Universal Mains Power Supply
Due to its low RDS(on), the NCP1027 can be used in
universal mains SMPS up to 15 W of continuous power,
provided that the chip power dissipation is well under
control. That is to say that average power calculations and
measurements have been carried and correlated. The
design of an SMPS around a monolithic device does not
differ from that of a standard circuit using a controller and
a MOSFET. However, one needs to be aware of certain
characteristics specific of monolithic devices. Let us
follow the steps:
Vin min = 120 Vdc
Vin max = 375 Vdc
Vout = 5.0 V
Vout = 15 W
Operating mode is CCM
h = 0.8
1. The lateral MOSFET body- diode shall never be
forward biased, either during startup (because of a
large leakage inductance) or in normal operation
as shown by Figure 45. This condition sets the
maximum voltage that can be reflected during toff .
1.004M 1.011M 1.018M 1.025M 1.032M
-50.0
50.0
150
250
350
> 0 !!
Figure 45. The reflected voltage shall always be greater
than the minimum input voltage to avoid the forward
biasing of the MOSFET body-diode.
Figure 46. Primary Inductance Current
Evolution in CCM
As a result, the Flyback voltage which is reflected on the
drain at the switch opening cannot be larger than the input
voltage. When selecting components, you thus must adopt
a turn ratio which adheres to the following equation:
N(Vout )Vf)tVin, min tVin min (eq. 14) . In our case,
since we operate from a 120 V DC rail while delivering
5.0 V, we can select a reflected voltage of 110 V
DC maximum: 120-110 > 0. Therefore, the turn ratio
Np:Ns must be smaller than Vin
Vout )Vf+110
5)1+18.3 or
Np : Ns t19. We will see later on how it affects the
calculation.
2. Lateral MOSFETs have a poorly doped
body-diode which naturally limits their ability to
sustain the avalanche. A traditional RCD
clamping network shall thus be installed to
protect the MOSFET. In some low power
applications, a simple capacitor can also be used
since Vdrain max +Vin )N(Vout )Vf)
)Ipeak Lf
Ctot
Ǹ(eq. 15) , where Lf is the leakage
inductance, Ctot the total capacitance at the drain
node (which is increased by the capacitor you
will wire between drain and source), N the Np:Ns
turn ratio, V
out the output voltage, Vf the
secondary diode forward drop and finally, Ipeak
the maximum peak current. Worse case occurs
when the SMPS is very close to regulation, e.g.
the V
out target is almost reached and Ipeak is still
pushed to the maximum. For this design, we have
selected our maximum voltage around 650 V (at
V
in = 375 Vdc). This voltage is given by the RCD
clamp installed from the drain to the bulk
voltage. We will see how to calculate it later on.
3. Calculate the maximum operating duty-cycle for
this flyback converter operated in CCM:
dmax +NVout
NVout )Vin, min +1
1)Vin,min
NVout
+0.49 (eq. 16)
4. To obtain the primary inductance, we have the
choice between two equations:
L+(Vind)2
fSWKPin
(eq. 17) , where K+DIL
I1 and
defines the amount of ripple we want in CCM
(see Figure 46).
•Small K: deep CCM, implying a large primary
inductance, a low bandwidth and a large leakage
inductance.
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•Large K: approaching BCM where the rms losses are
the worse, but smaller inductance, leading to a better
leakage inductance.
From Equation 16, a K factor of 0.8 (40% ripple), gives
an inductance of:
L+(120 0.49)2
60k 0.8 18.75 +3.8mH
DIL+Vind
LFSW +120 0.49
3.8m 60k +258mApeak *to *peak
The peak current can be evaluated to be:
Ipeak +Iavg
d)DIL
2+Ipeak +156m
0.49 )DIL
2+447mA
In Figure 46, I1 can also be calculated:
I1+Ipeak-DIL
2+0.447- 0.129 +318mA
5. Based on the above numbers, we can now
evaluate the conduction losses:
Id, rms +I1d
Ǹ1)1
3ǒDIL
2I1Ǔ2
Ǹ+0.318 0.7
1)1
3ǒ0.258
2 0.318Ǔ2
Ǹ+228mArms
If we take the maximum RDS(on) for a 120°C junction
temperature, i.e. 11 W, then conduction losses worse case
are:
Pcond +I2d, rmsRds(on) +571mW
6. Off-time and on-time switching losses can be
estimated based on the following calculations:
Poff +IpeakVdstoff
6TSW +0.447 650 40n
6 15u +130mW
(eq. 18)
Pon +IpeakN(Vout )Vf)ton
6TSW (eq. 19)
+0.447 114 40n
6 15u +22mW
The theoretical total power is then 0.571 + 0.13 + 0.022
= 723 mW.
7. The ramp compensation will be calculated as
suggested by Equation 13 giving a resistor of
78 kW or 82 kW for the normalized value.
Power Switch Circuit Protection
As in any Flyback design, it is important to limit the drain
excursion to a safe value, e.g. below the power switch
circuit BVdss which is 700 V. Figures 47a, b, c present
possible implementations:
+
1
2
3
45
7
8
Figure 47. Different Options to Clamp the Leakage Spike
6
CVCC
C
HV
+
1
2
3
45
7
8
6
CVCC
HV
Rclamp Cclamp
D
+
1
2
3
45
7
8
6
CVCC
HV
D
Dz
a. b. c.
Figure 47a: The simple capacitor limits the voltage
according to Equation 14. This option is only valid for low
power applications, e.g. below 5.0 W, otherwise chances
exist to destroy the MOSFET. After evaluating the leakage
inductance, you can compute C with Equation 15. Typical
values are between 100 pF and up to 470 pF. Large
capacitors increase capacitive losses…
Figure 47b: The most standard circuitry called the RCD
network. You calculate Rclamp and Cclamp using the
following formulae:
Rclamp +2Vclamp(Vclamp-(V
out )Vf)N)
LpeakI2
peak FSW
(eq. 20)
Cclamp +Vclamp
VrippleFSWRclamp
(eq. 21)
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clamp is usually selected 50-80 V above the reflected
value N (Vout + Vf). The diode needs to be a fast one and
an MUR160 represents a good choice. One major
drawback of the RCD network lies in its dependency upon
the peak current. Worse case occurs when Ipeak and Vin are
maximum and Vout is close to reach the steady-state value.
Figure 47c: This option is probably the most expensive
of all three but it offers the best protection degree. If you
need a very precise clamping level, you must implement a
Zener diode or a TVS. There are little technology
differences behind a standard Zener diode and a TVS.
However, the die area is far bigger for a transient suppressor
than that of Zener. A 5.0 W Zener diode, like the 1N5388B,
will accept 180 W peak power if it lasts less than 8.3 ms.
If the peak current in the worse case (e.g. when the PWM
circuit maximum current limit works) multiplied by the
nominal Zener voltage exceeds these 180 W, then the diode
will be destroyed when the supply experiences overloads.
A transient suppressor like the P6KE200 still dissipates
5.0 W of continuous power, but is able to accept surges up
to 600 W @ 1.0 ms. Select the Zener or TVS clamping
level between 40 to 80 V above the reflected output voltage
when the supply is heavily loaded.
Power Dissipation and Heatsinking
The NCP1027 hosting a power switch circuit and a
controller, it is mandatory to properly manage the heat
generated by losses. If no precaution is taken, risks exist to
trigger the internal thermal shutdown (TSD). To help
dissipating the heat, the PCB designer must foresee large
copper areas around the PDI7 package. When surrounded
by a surface greater than 1.0 cm@ of 35 mm copper, it
becomes possible to drop the thermal resistance
junction-to-ambient, RqJA down to 75°C/W and thus
dissipate more power. The maximum power the device can
thus evacuate is: Pmax +Tjmax - Tamb max
RqJA
(eq. 22)
which gives around 930 mW for an ambient of 50°C and a
maximum junction of 120°C. The losses inherent to the
switch circuit RDS(on) can be theoretically evaluated, but
the final prototype evaluation must include board
measurements to confirm that the junction temperature
stays within safe limits. Figure 48 gives a possible layout
to help dropping the thermal resistance. When measured on
a 70 mm (2 oz.) copper thickness PCB, we obtained a
thermal resistance of 75°C/W.
Figure 48. A possible PCB arrangement to reduce the thermal resistance
junction-to-ambient.
When routing the printed circuit, it is important to keep
high impedance line very short, like the brown-out signal
and the OPP input if used.
Application Diagram
Figure 49 displays the final application schematic. The
output uses a TLV431 whose low bias current represents an
advantage for low standby power switch mode supplies.
The secondary side features an additional LC filter needed
to remove unwanted spikes, although less problematic than
in DCM operation. On the primary side, a resistive network
senses the input bulk voltage and prevents the controller
from turning on for input voltages below 100 Vdc. The
auxiliary winding delivers 20 V nominal and thus offers
comfortable margin when the converter enters standby. As
we do not use any OPP, pin 7 goes to ground and offers
extended possibility to layout more copper area.
NCP1027
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28
85-265 VAC
+
+
1
2
3
45
7
8
U2
NCP1027
C13
220 nF
Type = X2
C4
47 mF/
400 V
R1
2.8 M
R10
200 k
R3
18 k
C1
10 n
CVCC
47 mF
R2
78 k C12
100 p
R9
5.6 k
20 V
+C11
1 mF
D2
1N4637
R5
150 k
Type = 1W
C7
10 nF
Type = 400 V
+
C5
470
mF
+
C8
470
mF
+
C9
470
mF
D1
MBRD640CTT4
C10
2.2 nF
Type = Y1
R4
100
U1
TL431
R11
1 k
R6
10 k
+
C3
100 mF
C2
100 nF
R7
10 k
Vout
5 V @
3 A
L2
2.2 mH
Np:Ns = 1:0.062
Np:Naux = 1:0.208
Lp = 3.8 mH
Figure 49. 5.0 V-3.0 A Universal Mains Power Supply
D5
1N4637
NCP1027
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29
Transformer Specifications:
Vout = 5.0 V/3.0 A
Vaux = 20 V/10 mA
Lp = 3.8 mH
Ip, rms = 280 mA
Ip, max = 800 mA
Isec, rms = 5.0 A
Fsw = 65 kHz
Np:Nsec = 1 : 0.052
Np:Naux = 1 : 0.208
NCP1027
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30
PACKAGE DIMENSIONS
8-LEAD PDIP
P SUFFIX
CASE 626A-01
ISSUE O
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. PACKAGE CONTOUR OPTIONAL (ROUND OR
SQUARE CORNERS).
4. DIMENSION L TO CENTER OF LEAD WHEN
FORMED PARALLEL.
5. DIMENSIONS A AND B ARE DATUMS.
14
58
F
NOTE 3
-T-
SEATING
PLANE
H
J
G
DK
N
C
L
M
M
A
M
0.13 (0.005) B M
T
DIM MIN MAX MIN MAX
INCHESMILLIMETERS
A9.40 10.16 0.370 0.400
B6.10 6.60 0.240 0.260
C3.94 4.45 0.155 0.175
D0.38 0.51 0.015 0.020
F1.02 1.78 0.040 0.070
G2.54 BSC 0.100 BSC
H0.76 1.27 0.030 0.050
J0.20 0.30 0.008 0.012
K2.92 3.43 0.115 0.135
L7.62 BSC 0.300 BSC
M--- 10 --- 10
N0.76 1.01 0.030 0.040
__
B
A
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NCP1027D
The product described herein (NCP1027), may be covered by one or more of the following U.S. patents: 5,418,410; 5,477,175; 6,271,735;
6,362,067; 6,385,060; 6,429,709; 6,448,625; 6,492,679; 6,597,221; 6,633,193; 6,919,598; 6,940,320. There may be other patents pending.
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