EVALUATION KIT AVAILABLE MAX16932/MAX16933 General Description The MAX16932/MAX16933 offer two high-voltage, synchronous step-down controllers that use only 20A of quiescent current with no load. They operate with an input voltage supply from 3.5V to 42V and can operate in dropout condition by running at 95% duty cycle. The devices are intended for applications with mid- to high-power requirements and requiring two independently controlled output supplies, such as automotive applications. The MAX16932/MAX16933 step-down controllers operate 180 out-of-phase for reduced input ripple. The devices also operate with switching frequencies up to 2.2MHz to allow use of small external components and to guarantee no AM band interference. The FSYNC input programmability enables three frequency modes for optimized performance: forced fixed-frequency operation, skip mode with ultra-low quiescent current (20A), and synchronization to an external clock. The devices provide a spread-spectrum option to minimize EMI interference. 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Benefits and Features Meets Stringent OEM Module Power Consumption and Performance Specifications * 20A Quiescent Current in Skip Mode * 1% Output-Voltage Accuracy: 5.0V/3.3V Fixed or Adjustable Between 1V and 10V Enables Crank-Ready Designs * Wide Input Supply Range from 3.5V to 36V EMI Reduction Features Reduce Interference with Sensitive Radio Bands without Sacrificing Wide Input Voltage Range * 50ns (typ) Minimum On-Time Guarantees SkipFree Operation for 3.3V Output from Car Battery at 2.2MHz * Spread-Spectrum Option * Frequency-Synchronization Input * Resistor-Programmable Frequency Between 200kHz and 2.2MHz The devices are available in a 28-pin TQFN-EP package and are specified for operation over the -40C to +125C automotive temperature range. Integration and Thermally Enhanced Packages Save Board Space and Cost * Dual, 2MHz Step-Down Controllers * 180 Out-of-Phase Operation * Current-Mode Controllers with Forced-Continuous and Skip Modes * Thermally Enhanced 28-Pin TQFN-EP Package Ordering Information and Selector Guide appears at end of data sheet. Protection Features Improve System Reliability * Supply Overvoltage and Undervoltage Lockout * Overtemperature and Short-Circuit Protection The devices also feature a power-OK monitor and overvoltage and undervoltage lockout. Protection features include cycle-by-cycle current limit and thermal shutdown. Applications POL Applications for Automotive Power Distributed DC Power Systems Navigation and Radio Head Units 19-6716; Rev 10; 9/17 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Absolute Maximum Ratings IN, EN1, EN2, TERM to PGND_............................-0.3V to +42V CS1, CS2, OUT1, OUT2 to AGND......................... -0.3V to +11V CS1 to OUT1.........................................................-0.2V to +0.2V CS2 to OUT2.........................................................-0.2V to +0.2V BIAS, FSYNC, FOSC to AGND............................-0.3V to +6.0V COMP1, COMP2 to AGND...................................-0.3V to +6.0V FB1, FB2, EXTVCC to AGND...............................-0.3V to +6.0V DL_ to PGND_......................................................-0.3V to +6.0V BST_, to LX_.........................................................-0.3V to +6.0V DH_ to LX_............................................................-0.3V to +6.0V LX_ to PGND_.......................................................-0.3V to +42V PGND_ to AGND...................................................-0.3V to +0.3V PGOOD1, PGOOD2 to AGND..............................-0.3V to +6.0V Continuous Power Dissipation (TA = +70C) TQFN (derate 28.6mW/NC above +70C)..............2285.7mW Operating Temperature Range.......................... -40C to +125C Junction Temperature Range...........................................+150C Storage Temperature Range............................. -65C to +150C Lead Temperature (soldering, 10s).................................. +300C Soldering Temperature (reflow)........................................ +260C Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Package Thermal Characteristics (Note 1) TQFN Junction-to-Ambient Thermal Resistance (JA)...........35C/W Junction-to-Case Thermal Resistance (JC)..................3C/W Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial. Electrical Characteristics (VIN = 14V, VBIAS = 5V, CBIAS = 6.8F, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C under normal conditions, unless otherwise noted.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNIT SYNCHRONOUS STEP-DOWN DC-DC CONTROLLERS Supply Voltage Range VIN Output Overvoltage Threshold Supply Current IIN Buck 1 Fixed Output Voltage VOUT1 Buck 2 Fixed Output Voltage VOUT2 Output Voltage Adjustable Range www.maximintegrated.com Normal operation 3.5 36 t < 1s 42 FB rising (Note 3) +10 +15 +20 FB falling +5 +10 +15 VEN1 = VEN2 = 0V, TA = +25C 8 20 VEN1 = VEN2 = 0V, TA = +125C 20 VEN1 = 5V, VOUT1 = 5V, VEN2 = 0V; VEXTVCC = 5V, no switching 30 40 VEN2 = 5V, VOUT2 = 3.3V; VEN1 = 0V, VEXTVCC = 3.3V, no switching 20 30 VEN1 = VEN2 = 5V, VOUT1 = 5V, VOUT2 = 3.3V, VEXTVCC = 3.3V, no switching 25 40 VFB1 = VBIAS, PWM mode 4.95 5 5.05 VFB1 = VBIAS, skip mode 4.95 5 5.075 VFB2 = VBIAS, PWM mode 3.234 3.3 3.366 VFB2 = VBIAS, skip mode 3.234 3.3 3.4 Buck 1, buck 2 1 10 V % A V V V Maxim Integrated 2 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Electrical Characteristics (continued) (VIN = 14V, VBIAS = 5V, CBIAS = 6.8F, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C under normal conditions, unless otherwise noted.) (Note 2) PARAMETER SYMBOL Regulated Feedback Voltage VFB1,2 Feedback Leakage Current IFB1,2 Feedback Line Regulation Error Transconductance (from FB_ to COMP_) gm Dead Time Maximum Duty Cycle Minimum On-Time PWM Switching Frequency CONDITIONS 0.99 fSW Switching Frequency Accuracy Spread-Spectrum Range MAX UNIT 1.0 1.01 V 0.01 1 A VIN = 3.5V to 36V, VFB = 1V 0.001 VFB = 1V, VBIAS = 5V 1200 MAX16932: DL_ low to DH_ high 35 MAX16932: DH_ low to DL_ high 60 MAX16933: DL_ low to DH_ high 60 MAX16933: DH_ low to DL_ high 100 95 Buck 1, buck 2 %/V 2400 S ns 98.5 % 50 ns MAX16932 1 2.2 MAX16933 0.2 1 MAX16932ATIT/V+, MAX16932CATIU/V+ only Buck 2 Switching Frequency TYP TA = +25C Buck 1, buck 2 tON(MIN) MIN 1/2fSW MHz MHz MAX16932: RFOSC = 13.7k, VBIAS = 5V 1.98 2.2 2.42 MHz MAX16933: RFOSC = 80.6k, VBIAS = 5V 360 400 440 kHz Spread spectrum enabled 6 % FSYNC INPUT FSYNC Frequency Range FSYNC Switching Thresholds CS Current-Limit Voltage Threshold MAX16932: Minimum sync pulse of 100ns 1.2 2.4 MHz MAX16933: Minimum sync pulse of 400ns 240 1200 kHz High threshold 1.5 Low threshold VLIMIT1,2 VCS - VOUT, VBIAS = 5V, VOUT 2.5V 0.6 64 Skip Mode Threshold Soft-Start Ramp Time 80 96 15 Buck 1 and buck 2, fixed soft-start time regardless of frequency Phase Shift Between Buck1 and Buck 2 2 6 V mV mV 10 ms 180 0.01 A LX1, LX2 Leakage Current VIN = 6V, VLX_ = VIN, TA = +25C DH1, DH2 Pullup Resistance VBIAS = 5V, IDH_ = -100mA 10 20 DH1, DH2 Pulldown Resistance VBIAS = 5V, IDH_ = +100mA 2 4 DL1, DL2 Pullup Resistance VBIAS = 5V, IDL_ = -100mA 4 8 DL1, DL2 Pulldown Resistance VBIAS = 5V, IDL_ = +100mA 1.5 3 www.maximintegrated.com Maxim Integrated 3 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Electrical Characteristics (continued) (VIN = 14V, VBIAS = 5V, CBIAS = 6.8F, TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C under normal conditions, unless otherwise noted.) (Note 2) PARAMETER PGOOD1, PGOOD2 Threshold SYMBOL CONDITIONS MIN TYP MAX PGOOD_H % of VOUT_, rising 85 90 95 PGOOD_F % of VOUT_, falling 80 85 90 0.01 1 PGOOD1, PGOOD2 Leakage Current VPGOOD1,2 = 5V, TA = +25C PGOOD1, PGOOD2 Startup Delay Time Buck 1 and buck 2 after soft-start is complete PGOOD1, PGOOD2 Debounce Time Fault detection 64 UNIT % A Cycles 8 20 50 s 4.75 5 5.25 V 3.1 3.4 INTERNAL LDO: BIAS Internal BIAS Voltage VIN > 6V VBIAS rising BIAS UVLO Threshold VBIAS falling 2.7 Hysteresis External VCC 2.9 0.2 VTH,EXTVCC EXTVCC rising, HYST = 110mV 3.0 V V 3.2 V THERMAL OVERLOAD Thermal Shutdown Temperature (Note 4) +170 C Thermal Shutdown Hysteresis (Note 4) 20 C EN LOGIC INPUT High Threshold 1.8 Low Threshold Input Current TA = +25C V 0.8 V 1 A Note 2: Limits are 100% production tested at TA = +25C. Limits over the operating temperature range and relevant supply voltage range are guaranteed by design and characterization. Typical values are at TA = +25C. Note 3: Overvoltage protection is detected at the FB1/FB2 pins. If the feedback voltage reaches overvoltage threshold of FB1/FB2 + 15% (typ), the corresponding controller stops switching. The controllers resume switching once the output drops below FB1/ FB2 + 10% (typ). Note 4: Guaranteed by design; not production tested. www.maximintegrated.com Maxim Integrated 4 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Typical Operating Characteristics (TA = +25C, unless otherwise noted.) NO LOAD STARTUP SEQUENCE (VFSYNC = 0V) FULL LOAD STARTUP SEQUENCE (VFSYNC = 0V) MAX16932 toc01 MAX16932 toc02 VBAT 5V/div VBAT 5V/div VOUT1 2V/div IOUT1 2A/div VPGOOD1 5V/div VOUT1 2V/div VOUT2 2V/div VOUT2 2V/div IOUT2 2A/div VPGOOD2 5V/div VPGOOD1 5V/div VPGOOD2 5V/div QUIESCIENT CURRENT vs. SUPPLY VOLTAGE 30 20 VEN1 = 0V VEN2 = VBAT EXTVCC = VOUT2 100 90 EFFICIENCY (%) 80 70 60 50 BUCK1 EFFICIENCY 50 40 30 20 EXTVCC = VOUT2 EXTVCC = GND PWM MODE 10 0 1.0E-04 1.0E-02 1.0E+00 1.0E-06 1.0E-05 1.0E-03 1.0E-01 1.0E+01 IOUT1 (A) www.maximintegrated.com BUCK 1 EXTVCC = VOUT1 40 30 20 0 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) fSW = 2.2MHz EXTVCC = VOUT1 L = 2.2H VBAT = 14V VOUT1 = 5V EXTVCC = GND SKIP MODE MAX16932 toc04 60 BUCK 2 EXTVCC = VOUT2 10 MAX16932 toc05 0 70 100 90 80 70 60 0 5 10 15 20 25 30 SUPPLY VOLTAGE (V) 30 20 40 BUCK2 EFFICIENCY fSW = 2.2MHz EXTVCC = VOUT2 L = 2.2H VBAT = 14V VOUT2 = 3.3V EXTVCC = GND SKIP MODE 50 40 35 MAX16932 toc06 VEN1 = VBAT VEN2 = 0V EXTVCC = VOUT1 10 80 SUPPLY CURRENT (A) 40 QUIESCIENT CURRENT vs. TEMPERATURE EFFICIENCY (%) SUPPLY CURRENT (A) 50 4ms/div MAX16932 toc03 60 2ms/div EXTVCC = GND EXTVCC = VOUT2 PWM MODE 10 0 1.0E-04 1.0E-02 1.0E+00 1.0E-06 1.0E-05 1.0E-03 1.0E-01 1.0E+01 IOUT2 (A) Maxim Integrated 5 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Typical Operating Characteristics (continued) (TA = +25C, unless otherwise noted.) 2.22 BUCK 1 2.20 2.18 2.16 2.14 2.12 2.10 0 1 2 3 4 5 2.0 1.8 VBIAS = 5V 1.6 1.4 VBIAS = 3.3V 1.2 15 0 LOAD CURRENT (A) 0.9 0.8 0.7 0.6 VBIAS = 5V 0.5 0.4 VBIAS = 3.3V 25 30 0.2 30 40 50 60 RFOSC = 13.7k 2.30 70 90 110 130 150 170 80 100 120 140 160 RFOSC (k) LOAD TRANSIENT RESPONSE MAX16932 toc11 MAX16932 toc10 SWITCHING FREQUENCY (MHz) 2.35 20 RFOSC (k) SWITCHING FREQUENCY vs. TEMPERATURE 2.40 1.0 0.3 0 6 MAX16932 toc09 2.2 SWITCHING FREQUENCY vs. RFOSC (MAX16933) 1.1 SWITCHING FREQUENCY (MHz) 2.24 MAX16932 toc08 BUCK 2 2.26 2.4 SWITCHING FREQUENCY (MHz) 2.28 MAX16932 toc07 2.30 SWITCHING FREQUENCY (MHz) SWITCHING FREQUENCY vs. RFOSC (MAX16932) SWITCHING FREQUENCY vs. LOAD CURRENT VOUT1 100mV/div 2.25 2.20 2.15 2.10 IOUT1 1A/div 2.05 2.00 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) 400s/div EXTERNAL SYNC TRANSITION DIPS AND DROPS MAX16932 toc12 VLX1 10V/div VBAT 10V/div VLX2 10V/div VPGOOD1 5V/div VSYNC 2V/div 400ns/div www.maximintegrated.com MAX16932 toc13 VOUT1 5V/div 40ms/div Maxim Integrated 6 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Typical Operating Characteristics (continued) (TA = +25C, unless otherwise noted.) LOAD DUMP SLOW VIN RAMP MAX16932 toc14 MAX16932 toc15 VBAT 5V/div VBAT 10V/div VOUT2 1V/div VOUT1 2V/div VPGOOD1 5V/div VOUT2 2V/div VPGOOD2 5V/div VPGOOD2 5V/div 100ms/div 10s/div OUTPUT OVERVOLTAGE RESPONSE SHORT CIRCUIT RESPONSE MAX16932 toc17 MAX16932 toc16 VOUT1 1V/div VPGOOD1 2V/div IOUT1 2A/div VOUT1 1V/div VPGOOD1 2V/div 200s/div 1s/div 3.295 VOUT (V) VOUT (V) 4.995 4.994 4.993 4.992 4.991 4.990 4.989 VSYNC = VBIAS 0 1 2 3 IOUT (A) www.maximintegrated.com 4 5 6 100.05 100.00 VOUT (%NOMINAL) 4.996 VSYNC = VBIAS 3.296 100.10 3.294 3.293 3.292 99.95 3.290 99.75 1 2 3 IOUT (A) 4 5 6 VOUT1 99.85 99.80 0 VOUT2 99.90 3.291 3.289 MAX16932 toc20 MAX16932 toc18 VSYNC = VBIAS 4.997 VOUT vs. TEMPERATURE BUCK 2 LOAD REGULATION 3.297 MAX16932 toc19 BUCK 1 LOAD REGULATION 4.998 99.70 EXTVCC = VGND VSYNC = VBIAS IOUT_ = 0A -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) Maxim Integrated 7 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Typical Operating Characteristics (continued) (TA = +25C, unless otherwise noted.) MAX16932 toc21 VOUT1 = 1.8V 1.000 0.995 1.000 0.995 0 10 5 15 20 25 VSUP (V) 30 35 0.990 40 0 MINIMUM ON-TIME (BUCK 1) 10 5 15 20 25 VSUP (V) IOUT2 = 300mA VBAT 5V/div VBAT 5V/div VOUT1 1V/div VOUT1 1V/div 200ns/div 20 10 0 30 MEASURED AT VOUT2 ON THE MAX16932CATIU/V+ 25 20 15 10 5 0 500,000 -10 800k 30 SPECTRAL ENERGY DENSITY vs. FREQUENCY MEASURED ON THE MAX16932CATIS/V+ 25 20 15 10 5 0 -5 -5 350,000 400,000 450,000 FREQUENCY (Hz) 35 OUTPUT SPECTRUM (dBV) 30 35 SPECTRAL ENERGY DENSITY vs. FREQUENCY MAX16932 toc26 40 40 OUTPUT SPECTRUM (dBV) MEASURED ON THE MAX16933CATIS/V+ MAX16932 toc25 OUTPUT SPECTRUM (dBV) 200ns/div SPECTRAL ENERGY DENSITY vs. FREQUENCY www.maximintegrated.com 40 MAX16932 toc24 IOUT1 = 300mA -10 300,000 35 MINIMUM ON-TIME (BUCK 2) MAX16932 toc23 50 30 MAX16932 toc27 0.990 VOUT2 = 1.8V 1.005 VOUT (V) VOUT (V) 1.005 FB2 LINE REGULATION 1.010 MAX16932 toc22 FB1 LINE REGULATION 1.010 900k 1000k 1100k FREQUENCY (Hz) 1200k -10 1800k 2200k 2400k 2000k FREQUENCY (Hz) 2600k Maxim Integrated 8 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current FB2 COMP2 FOSC 20 CS2 21 OUT2 PGND2 TOP VIEW DL2 Pin Configuration 19 18 17 16 15 LX2 22 14 FSYNC DH2 23 13 PGOOD2 12 PGOOD1 11 IN EN1 26 10 EXTVCC BST1 27 9 AGND 8 BIAS MAX16932 MAX16933 BST2 24 EN2 25 EP 5 6 7 COMP1 4 FB1 LX1 DL1 3 OUT1 2 CS1 1 PGND1 + DH1 28 TQFN (5mm x 5mm) Pin Description PIN NAME 1 LX1 Inductor Connection for Buck 1. Connect LX1 to the switched side of the inductor. LX1 serves as the lower supply rail for the DH1 high-side gate drive. 2 DL1 Low-Side Gate Drive Output for Buck 1. DL1 output voltage swings from VPGND1 to VBIAS. 3 PGND1 4 CS1 5 OUT1 6 FB1 7 COMP1 8 BIAS 9 AGND www.maximintegrated.com DESCRIPTION Power Ground for Buck 1 Positive Current-Sense Input for Buck 1. Connect CS1 to the positive terminal of the current-sense resistor. See the Current Limiting and Current-Sense Inputs and Current-Sense Measurement sections. Output Sense and Negative Current-Sense Input for Buck 1. When using the internal preset 5V feedback divider (FB1 = BIAS), the buck uses OUT1 to sense the output voltage. Connect OUT1 to the negative terminal of the current-sense resistor. See the Current Limiting and Current-Sense Inputs and Current-Sense Measurement sections. Feedback Input for Buck 1. Connect FB1 to BIAS for the 5V fixed output or to a resistive divider between OUT1 and GND to adjust the output voltage between 1V and 10V. In adjustable mode, FB1 regulates to 1V (typ). See the Setting the Output Voltage in Buck Converters section. Buck 1 Error-Amplifier Output. Connect an RC network to COMP1 to compensate buck 1. 5V Internal Linear Regulator Output. Bypass BIAS to GND with a low-ESR ceramic capacitor of 6.8F minimum value. BIAS provides the power to the internal circuitry and external loads. See the Fixed 5V Linear Regulator (BIAS) section. Signal Ground for IC Maxim Integrated 9 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Pin Description (continued) PIN NAME 10 EXTVCC 11 IN Supply Input. Bypass IN with sufficient capacitance to supply the two out-of-phase buck converters. PGOOD1 Open-Drain Power-Good Output for Buck 1. PGOOD1 is low if OUT1 is more than 15% (typ) below the normal regulation point. PGOOD1 asserts low during soft-start and in shutdown. PGOOD1 becomes high impedance when OUT1 is in regulation. To obtain a logic signal, pullup PGOOD1 with an external resistor connected to a positive voltage lower than 5.5V. Place a minimum of 100 (RPGOOD1) in series with PGOOD1. See the Voltage Monitoring section for details. 13 PGOOD2 Open-Drain Power-Good Output for Buck 2. PGOOD2 is low if OUT2 is more than 15% (typ) below the normal regulation point. PGOOD2 asserts low during soft-start and in shutdown. PGOOD2 becomes high impedance when OUT2 is in regulation. To obtain a logic signal, pullup PGOOD2 with an external resistor connected to a positive voltage lower than 5.5V. 14 FSYNC External Clock Synchronization Input. Synchronization to the controller operating frequency ratio is 1. Keep fSYNC a minimum of 10% greater than the maximum internal switching frequency for stable operation. See the Switching Frequency/External Synchronization section. 15 FOSC Frequency Setting Input. Connect a resistor from FOSC to AGND to set the switching frequency of the DC-DC converters. 16 COMP2 17 FB2 12 DESCRIPTION 3.1V to 5.2V Input to the Switchover Comparator Buck 2 Error Amplifier Output. Connect an RC network to COMP2 to compensate buck 2. Feedback Input for Buck 2. Connect FB2 to BIAS for the 3.3V fixed output or to a resistive divider between OUT2 and GND to adjust the output voltage between 1V and 10V. In adjustable mode, FB2 regulates to 1V (typ). See the Setting the Output Voltage in Buck Converters section. Output Sense and Negative Current-Sense Input for Buck 2. When using the internal preset 3.3V feedback-divider (FB2 = BIAS), the buck uses OUT2 to sense the output voltage. Connect OUT2 to the negative terminal of the current-sense resistor. See the Current Limiting and Current-Sense Inputs and Current-Sense Measurement sections. 18 OUT2 19 CS2 20 PGND2 21 DL2 Low-Side Gate Drive Output for Buck 2. DL2 output voltage swings from VPGND2 to VBIAS. 22 LX2 Inductor Connection for Buck 2. Connect LX2 to the switched side of the inductor. LX2 serves as the lower supply rail for the DH2 high-side gate drive. 23 DH2 High-Side Gate Drive Output for Buck 2. DH2 output voltage swings from VLX2 to VBST2. 24 BST2 Boost Capacitor Connection for High-Side Gate Voltage of Buck 2. Connect a high-voltage diode between BIAS and BST2. Connect a ceramic capacitor between BST2 and LX2. See the High-Side Gate-Driver Supply (BST_) section. 25 EN2 High-Voltage Tolerant, Active-High Digital Enable Input for Buck 2. Driving EN2 high enables buck 2. 26 EN1 High-Voltage Tolerant, Active-High Digital Enable Input for Buck 1. Driving EN1 high enables buck 1. www.maximintegrated.com Positive Current-Sense Input for Buck 2. Connect CS2 to the positive terminal of the current-sense resistor. See the Current Limiting and Current-Sense Inputs and Current-Sense Measurement sections. Power Ground for Buck 2 Maxim Integrated 10 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Pin Description (continued) PIN NAME 27 BST1 Boost Capacitor Connection for High-Side Gate Voltage of Buck 1. Connect a high-voltage diode between BIAS and BST1. Connect a ceramic capacitor between BST1 and LX1. See the High-Side Gate-Driver Supply (BST_) section. 28 DH1 High-Side Gate-Drive Output for Buck 1. DH1 output voltage swings from VLX1 to VBST1. -- EP DESCRIPTION Exposed Pad. Connect the exposed pad to ground. Connecting the exposed pad to ground does not remove the requirement for proper ground connections to PGND1, PGND2, and AGND. The exposed pad is attached with epoxy to the substrate of the die, making it an excellent path to remove heat from the IC. Detailed Description The MAX16932/MAX16933 are automotive-rated dualoutput switching power supplies. These devices integrate two synchronous step-down controllers and can provide two independent controlled power rails as follows: * A buck controller with a fixed 5V output voltage or an adjustable 1V to 10V output voltage. * A buck controller with a fixed 3.3V output voltage or an adjustable 1V to 10V output voltage. The two buck controllers can each provide up to 10A output current and are independently controllable. EN1 and EN2 enable the respective buck controllers. Connect EN1 and EN2 directly to VBAT, or to powersupply sequencing logic. In skip mode, with no load and only buck 2 active, the total supply current is reduced to 20A (typ). When both controllers are disabled, the total current drawn is further reduced to 8A (typ). Fixed 5V Linear Regulator (BIAS) The internal circuitry of the MAX16932/MAX16933 requires a 5V bias supply. An internal 5V linear regulator (BIAS) generates this bias supply. Bypass BIAS with a 6.8F or greater ceramic capacitor to guarantee stability under the full-load condition. The internal linear regulator can source up to 100mA (150mA under EXTVCC switchover, see the EXTVCC Switchover section). Use the following equation to estimate the internal current requirements for the MAX16932/ MAX16933: www.maximintegrated.com IBIAS = ICC + fSW(QG_DH1 + QG_DL1 + QG_DH2 + QG_DL2) = 10mA to 50mA (typ) where ICC is the internal supply current, 5mA (typ), fSW is the switching frequency, and QG_ is the MOSFET's total gate charge (specification limits at VGS = 5V). To minimize the internal power dissipation, bypass BIAS to an external 5V rail. EXTVCC Switchover The internal linear regulator can be bypassed by connecting an external supply (3V to 5.2V) or the output of one of the buck converters to EXTVCC. BIAS internally switches to EXTVCC and the internal linear regulator turns off. This configuration has several advantages: * It reduces the internal power dissipation of the MAX16932/MAX16933. * The low-load efficiency improves as the internal supply current gets scaled down proportionally to the duty cycle. If VEXTVCC drops below VTH,EXTVCC = 3V (min), the internal regulator enables and switches back to BIAS. Undervoltage Lockout (UVLO) The BIAS input undervoltage lockout (UVLO) circuitry inhibits switching if the 5V bias supply (BIAS) is below its 2.9V (typ) UVLO falling threshold. Once the 5V bias supply (BIAS) rises above its UVLO rising threshold and EN1 and EN2 enable the buck controllers, the controllers start switching and the output voltages begin to ramp up using soft-start. Maxim Integrated 11 MAX16932/MAX16933 Buck Controllers The MAX16932/MAX16933 provide two buck controllers with synchronous rectification. The step-down controllers use a PWM, current-mode control scheme. External logic-level MOSFETs allow for optimized load-current design. Fixed-frequency operation with optimal interleaving minimizes input ripple current from the minimum to the maximum input voltages. Output-current sensing provides an accurate current limit with a sense resistor or power dissipation can be reduced using lossless current sensing across the inductor. Soft-Start Once a buck converter is enabled by driving the corresponding EN_ high, the soft-start circuitry gradually ramps up the reference voltage during soft-start time (tSSTART = 6ms (typ)) to reduce the input surge currents during startup. Before the device can begin the soft-start, the following conditions must be met: 1) VBIAS exceeds the 3.4V (max) undervoltage-lockout threshold. 2) VEN_ is logic-high. Switching Frequency/External Synchronization The MAX16932 provides an internal oscillator adjustable from 1MHz to 2.2MHz. The MAX16933 provides an internal oscillator adjustable from 200kHz to 1MHz. High-frequency operation optimizes the application for the smallest component size, trading off efficiency to higher switching losses. Low-frequency operation offers the best overall efficiency at the expense of component size and board space. To set the switching frequency, connect a resistor RFOSC from FOSC to AGND. See TOC8 and TOC9 (Switching Frequency vs. RFOSC) in the Typical Operating Characteristics to determine the relationship between switching frequency and RFOSC. Buck 1 is synchronized with the internal clock-signal rising edge, while buck 2 is synchronized with the clock-signal falling edge. The devices can be synchronized to an external clock by connecting the external clock signal to FSYNC. A rising edge on FSYNC resets the internal clock. Keep the FSYNC frequency between 110% and 150% of the internal frequency. The FSYNC signal should have a 50% duty cycle. www.maximintegrated.com 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Light-Load Efficiency Skip Mode (VFSYNC = 0V) Drive FSYNC low to enable skip mode. In skip mode, the devices stop switching until the FB voltage drops below the reference voltage. Once the FB voltage has dropped below the reference voltage, the devices begin switching until the inductor current reaches 20% (skip threshold) of the maximum current defined by the inductor DCR or output shunt resistor. Forced-PWM Mode (VFSYNC = High) Driving FSYNC high prevents the devices from entering skip mode by disabling the zero-crossing detection of the inductor current. This forces the low-side gate-driver waveform to constantly be the complement of the highside gate-drive waveform, so the inductor current reverses at light loads and discharges the output capacitor. The benefit of forced PWM mode is to keep the switching frequency constant under all load conditions. However, forced-frequency operation diverts a considerable amount of the output current to PGND, reducing the efficiency under light-load conditions. Forced-PWM mode is useful for improving load-transient response and eliminating unknown frequency harmonics that may interfere with AM radio bands. Spread Spectrum The MAX16932CATIS, MAX16932CATIU, and MAX16933CATIS feature enhanced EMI performance. They perform 6% dithering of the switching frequency to reduce peak emission noise at the clock frequency and its harmonics, making it easier to meet stringent emission limits. When using an external clock source (i.e., driving the FSYNC input with an external clock), spread spectrum is disabled. Buck 2 Switching Frequency For the MAX16932ATIT and MAX16932CATIU, the switching frequency of buck 2 is set to 1/2 of fSW (buck 1 switching frequency). When using these devices, the external components of buck 2 should be sized to account for the reduced switching frequency (see the Design Procedure section). Maxim Integrated 12 MAX16932/MAX16933 MOSFET Gate Drivers (DH_ and DL_) The DH_ high-side n-channel MOSFET drivers are powered from capacitors at BST_ while the low-side drivers (DL_) are powered by the 5V linear regulator (BIAS). On each channel, a shoot-through protection circuit monitors the gate-to-source voltage of the external MOSFETs to prevent a MOSFET from turning on until the complementary switch is fully off. There must be a low-resistance, low-inductance path from the DL_ and DH_ drivers to the MOSFET gates for the protection circuits to work properly. Follow the instructions listed to provide the necessary lowresistance and low-inductance path: 2.2MHz, 36V, Dual Buck with 20A Quiescent Current The boost capacitor should be a low-ESR ceramic capacitor. A minimum value of 100nF works in most cases. Current Limiting and Current-Sense Inputs (OUT_ and CS_) It may be necessary to decrease the slew rate for the gate drivers to reduce switching noise or to compensate for low-gate charge capacitors. For the low-side drivers, use gate capacitors in the range of 1nF to 5nF from DL_ to GND. For the high-side drivers, connect a small 5 to 10 resistor between BST_ and the bootstrap capacitor. Note: Gate drivers must be protected during shutdown, at the absence of the supply voltage (VBIAS = 0V) when the gate is pulled high either capacitively or by the leakage path on the PCB. Therefore, external gate pulldown resistors are needed, to prevent making a direct path from VBAT to GND. The current-limit circuit uses differential current-sense inputs (OUT_ and CS_) to limit the peak inductor current. If the magnitude of the current-sense signal exceeds the current-limit threshold (VLIMIT1,2 = 80mV (typ)), the PWM controller turns off the high-side MOSFET. The actual maximum load current is less than the peak current-limit threshold by an amount equal to half of the inductor ripple current. Therefore, the maximum load capability is a function of the current-sense resistance, inductor value, switching frequency, and duty cycle (VOUT_/VIN). For the most accurate current sensing, use a currentsense shunt resistor (RSH) between the inductor and the output capacitor. Connect CS_ to the inductor side of RSH and OUT_ to the capacitor side. Dimension RSH such that the maximum inductor current (IL,MAX = ILOAD,MAX+1/2 IRIPPLE,PP) induces a voltage of VLIMIT1,2 across RSH including all tolerances. For higher efficiency, the current can also be measured directly across the inductor. This method could cause up to 30% error over the entire temperature range and requires a filter network in the current-sense circuit. See the Current-Sense Measurement section. High-Side Gate-Driver Supply (BST_) Voltage Monitoring (PGOOD_) * Use very short, wide traces (50 mils to 100 mils wide if the MOSFET is 1in from the driver). The high-side MOSFET is turned on by closing an internal switch between BST_ and DH_ and transferring the bootstrap capacitor's (at BST_) charge to the gate of the high-side MOSFET. This charge refreshes when the highside MOSFET turns off and the LX_ voltage drops down to ground potential, taking the negative terminal of the capacitor to the same potential. At this time the bootstrap diode recharges the positive terminal of the bootstrap capacitor. The selected n-channel high-side MOSFET determines the appropriate boost capacitance values (CBST_ in the Typical Operating Circuit) according to the following equation: C BST_ = QG VBST_ where QG is the total gate charge of the high-side MOSFET and VBST_ is the voltage variation allowed on the high-side MOSFET driver after turn-on. Choose VBST_ such that the available gate-drive voltage is not significantly degraded (e.g., VBST_ = 100mV to 300mV) when determining CBST_. www.maximintegrated.com The MAX16932/MAX16933 include several power monitoring signals to facilitate power-supply sequencing and supervision. PGOOD_ can be used to enable circuits that are supplied by the corresponding voltage rail, or to turn on subsequent supplies. Each PGOOD_ goes high (high impedance) when the corresponding regulator output voltage is in regulation. Each PGOOD_ goes low when the corresponding regulator output voltage drops below 15% (typ) or rises above 15% (typ) of its nominal regulated voltage. Connect a 10k (typ) pullup resistor from PGOOD_ to the relevant logic rail to level-shift the signal. PGOOD_ asserts low during soft-start, soft-discharge, and when either buck converter is disabled (either EN1 or EN2 is low). To ensure latchup immunity on the PGOOD1 pin, in compliance with the AEC-Q100 guidelines, a minimum resistance of 100 should be placed between the PGOOD1 pin and any other external components. All other pins are compliant with no additional external components. Maxim Integrated 13 MAX16932/MAX16933 Thermal-Overload, Overcurrent, and Overvoltage and Undervoltage Behavior Thermal-Overload Protection Thermal-overload protection limits total power dissipation in the devices. When the junction temperature exceeds +170C, an internal thermal sensor shuts down the devices, allowing them to cool. The thermal sensor turns on the devices again after the junction temperature cools by 20C. Overcurrent Protection If the inductor current in the MAX16932/MAX16933 exceeds the maximum current limit programmed at CS_ and OUT_, the respective driver turns off. In an overcurrent mode, this results in shorter and shorter high-side pulses. A hard short results in a minimum on-time pulse every clock cycle. Choose the components so they can withstand the short-circuit current if required. Overvoltage Protection The devices limit the output voltage of the buck converters by turning off the high-side gate driver at approximately 115% of the regulated output voltage. The output voltage needs to come back in regulation before the high-side gate driver starts switching again. Design Procedure Buck Converter Design Procedure Effective Input Voltage Range in Buck Converters Although the MAX16932/MAX16933 can operate from input supplies up to 36V (42V transients) and regulate down to 1V, the minimum voltage conversion ratio (VOUT/VIN) might be limited by the minimum controllable on-time. For proper fixed-frequency PWM operation and optimal efficiency, buck 1 and buck 2 should operate in continuous conduction during normal operating conditions. For continuous conduction, set the voltage conversion ratio as follows: VOUT > t ON(MIN) x f SW VIN 2.2MHz, 36V, Dual Buck with 20A Quiescent Current where tON(MIN) is 50ns (typ) and fSW is the switching frequency in Hz. If the desired voltage conversion does not meet the above condition, pulse skipping occurs to decrease the effective duty cycle. Decrease the switching frequency if constant switching frequency is required. The same is true for the maximum voltage conversion ratio. The maximum voltage conversion ratio is limited by the maximum duty cycle (95%). VOUT < 0.95 VIN - VDROP where VDROP = IOUT (RON,HS + RDCR) is the sum of the parasitic voltage drops in the high-side path and fSW is the programmed switching frequency. During low drop operation, the devices reduce fSW to 25% (max) of the programmed frequency. In practice, the above condition should be met with adequate margin for good load-transient response. Setting the Output Voltage in Buck Converters Connect FB1 and FB2 to BIAS to enable the fixed buck controller output voltages (5V and 3.3V) set by a preset internal resistive voltage-divider connected between the output (OUT_) and AGND. To externally adjust the output voltage between 1V and 10V, connect a resistive divider from the output (OUT_) to FB_ to AGND (see the Typical Operating Circuit). Calculate RFB_1 and RFB_2 with the following equation: VOUT_ - 1 = R FB_1 R FB_2 VFB_ where VFB_ = 1V (typ) (see the Electrical Characteristics table). DC output accuracy specifications in the Electrical Characteristics table refer to the error comparator's threshold, VFB_ = 1V (typ). When the inductor conducts continuously, the devices regulate the peak of the output ripple, so the actual DC output voltage is lower than the slopecompensated trip level by 50% of the output ripple voltage. In discontinuous conduction mode (skip or STDBY active and IOUT < ILOAD(SKIP)), the devices regulate the valley of the output ripple, so the output voltage has a DC regulation level higher than the error-comparator threshold. www.maximintegrated.com Maxim Integrated 14 MAX16932/MAX16933 Inductor Selection in Buck Converters Three key inductor parameters must be specified for operation with the MAX16932/MAX16933: inductance value (L), inductor saturation current (ISAT), and DC resistance (RDCR). To determine the optimum inductance, knowing the typical duty cycle (D) is important. = D VOUT VOUT = OR D VIN VIN - I OUT (R DS(ON) + R DCR ) if the RDCR of the inductor and RDS(ON) of the MOSFET are available with VIN = (VBAT - VDIODE). All values should be typical to optimize the design for normal operation. Inductance The exact inductor value is not critical and can be adjusted in order to make trade-offs among size, cost, efficiency, and transient response requirements. * Lower inductor values increase LIR, which minimizes size and cost and improves transient response at the cost of reduced efficiency due to higher peak currents. * Higher inductance values decrease LIR, which increases efficiency by reducing the RMS current at the cost of requiring larger output capacitors to meet load-transient specifications. The ratio of the inductor peak-to-peak AC current to DC average current (LIR) must be selected first. A good initial value is a 30% peak-to-peak ripple current to averagecurrent ratio (LIR = 0.3). The switching frequency, input voltage, output voltage, and selected LIR then determine the inductor value as follows: L[H] = (VIN - VOUT )x D f SW [MHz]xI OUT x LIR where VIN, VOUT, and IOUT are typical values (so that efficiency is optimum for typical conditions). www.maximintegrated.com 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Peak Inductor Current Inductors are rated for maximum saturation current. The maximum inductor current equals the maximum load current in addition to half of the peak-to-peak ripple current: = IPEAK ILOAD(MAX) + IINDUCTOR 2 For the selected inductance value, the actual peak-to-peak inductor ripple current (IINDUCTOR) is calculated as: VOUT (VIN - VOUT ) IINDUCTOR = VIN x f SW x L where IINDUCTOR is in mA, L is in H, and fSW is in kHz. Once the peak current and the inductance are known, the inductor can be selected. The saturation current should be larger than IPEAK or at least in a range where the inductance does not degrade significantly. The MOSFETs are required to handle the same range of current without dissipating too much power. MOSFET Selection in Buck Converters Each step-down controller drives two external logic-level n-channel MOSFETs as the circuit switch elements. The key selection parameters to choose these MOSFETs include the items in the following sections. Threshold Voltage All four n-channel MOSFETs must be a logic-level type with guaranteed on-resistance specifications at VGS = 4.5V. If the internal regulator is bypassed (for example: VEXTVCC = 3.3V), then the n-channel MOSFETs should be chosen to have guaranteed on-resistance at that gateto-source voltage. Maximum Drain-to-Source Voltage (VDS(MAX)) All MOSFETs must be chosen with an appropriate VDS rating to handle all VIN voltage conditions. Maxim Integrated 15 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Current Capability sense resistor between the inductor and output as shown in Figure 1A. This configuration constantly monitors the inductor current, allowing accurate current-limit protection. Use low-inductance currentsense resistors for accurate measurement. The n-channel MOSFETs must deliver the average current to the load and the peak current during switching. Choose MOSFETs with the appropriate average current at VGS = 4.5V or VGS = VEXTVCC when the internal linear regulator is bypassed. For load currents below approximately 3A, dual MOSFETs in a single package can be an economical solution. To reduce switching noise for smaller MOSFETs, use a series resistor in the BST_ path and additional gate capacitance. Contact the factory for guidance using gate resistors. Alternatively, high-power applications that do not require highly accurate current-limit protection can reduce the overall power dissipation by connecting a series RC circuit across the inductor (Figure 1B) with an equivalent time constant: R2 R CSHL = R DCR R1 + R2 Current-Sense Measurement For the best current-sense accuracy and overcurrent protection, use a 1% tolerance current- INPUT (VIN) CIN MAX16932 MAX16933 DH_ NH DL_ RSENSE L LX_ COUT NL GND CS_ OUT_ A) OUTPUT SERIES RESISTOR SENSING INPUT (VIN) CIN MAX16932 MAX16933 DH_ NH LX_ DL_ NL GND CS_ OUT_ INDUCTOR L DCR R1 R2 CEQ COUT RCSHL = RDCR = ( ) [ ] R2 R R1 + R2 DCR L 1+ 1 CEQ R1 R2 B) LOSSLESS INDUCTOR SENSING Figure 1. Current-Sense Configurations www.maximintegrated.com Maxim Integrated 16 MAX16932/MAX16933 and: R DCR = L 1 1 + C EQ R1 R2 where RCSHL is the required current-sense resistor and RDCR is the inductor's series DC resistor. Use the inductance and RDCR values provided by the inductor manufacturer. Carefully observe the PCB layout guidelines to ensure the noise and DC errors do no corrupt the differential currentsense signals seen by CS_ and OUT_. Place the sense resistor close to the devices with short, direct traces, making a Kelvin-sense connection to the current-sense resistor. Input Capacitor in Buck Converters The discontinuous input current of the buck converter causes large input ripple currents and therefore the input capacitor must be carefully chosen to withstand the input ripple current and keep the input voltage ripple within design requirements. The 180 ripple phase operation increases the frequency of the input capacitor ripple current to twice the individual converter switching frequency. When using ripple phasing, the worst-case input capacitor ripple current is when the converter with the highest output current is on. The input voltage ripple is composed of VQ (caused by the capacitor discharge) and VESR (caused by the ESR of the input capacitor). The total voltage ripple is the sum of VQ and VESR that peaks at the end of an on-cycle. Calculate the input capacitance and ESR required for a specific ripple using the following equation: VESR ESR[] = IP - P ILOAD(MAX) + 2 V ILOAD(MAX) x OUT V IN C IN[F] = (VQ x f SW ) where: (VIN - VOUT ) x VOUT IP-P = VIN x f SW x L ILOAD(MAX) is the maximum output current in A, IP-P is the peak-to-peak inductor current in A, fSW is the switching frequency in MHz, and L is the inductor value in H. www.maximintegrated.com 2.2MHz, 36V, Dual Buck with 20A Quiescent Current The internal 5V linear regulator (BIAS) includes an output UVLO with hysteresis to avoid unintentional chattering during turn-on. Use additional bulk capacitance if the input source impedance is high. At lower input voltage, additional input capacitance helps avoid possible undershoot below the undervoltage lockout threshold during transient loading. Output Capacitor in Buck Converters The actual capacitance value required relates to the physical size needed to achieve low ESR, as well as to the chemistry of the capacitor technology. The capacitor is usually selected by ESR and the voltage rating rather than by capacitance value. When using low-capacity filter capacitors, such as ceramic capacitors, size is usually determined by the capacity needed to prevent VSAG and VSOAR from causing problems during load transients. Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem (see the Transient Considerations section). However, low-capacity filter capacitors typically have high-ESR zeros that can affect the overall stability. The total voltage sag (VSAG) can be calculated as follows: VSAG = L( ILOAD(MAX) ) 2 2C OUT ((VIN x D MAX ) - VOUT ) + ILOAD(MAX) (t - t) C OUT The amount of overshoot (VSOAR) during a full-load to no-load transient due to stored inductor energy can be calculated as: VSOAR ( ILOAD(MAX) ) 2 L 2C OUT VOUT ESR Considerations The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and load-transient requirements, yet have high enough ESR to satisfy stability requirements. When using high-capacitance, low-ESR capacitors, the filter capacitor's ESR dominates the output voltage ripple. So the output capacitor's size depends on the maximum ESR required to meet the output-voltage ripple (VRIPPLE(P-P)) specifications: VRIPPLE(P-P) = ESR x ILOAD(MAX) x LIR Maxim Integrated 17 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current In standby mode, the inductor current becomes discontinuous, with peak currents set by the idle-mode current-sense threshold (VCS,SKIP = 26mV (typ)). gmc = 1/(AVCS x RDC) CS_ CURRENT MODE POWER MODULATION Transient Considerations The output capacitor must be large enough to absorb the inductor energy while transitioning from no-load to full-load condition without tripping the overvoltage fault protection. The total output voltage sag is the sum of the voltage sag while the inductor is ramping up and the voltage sag before the next pulse can occur. Therefore: ( ) 2 L ILOAD(MAX) C OUT = 2VSAG (VIN x D MAX - VOUT ) + OUT_ R1 RESR COUT gmea = 1200S FB_ COMP_ ERROR AMP R2 VREF RC 30M CF CC ILOAD(MAX) (t - t) VSAG where DMAX is the maximum duty factor (approximately 95%), L is the inductor value in H, COUT is the output capacitor value in F, t is the switching period (1/fSW) in s, and t equals (VOUT/VIN) x t. The MAX16932/MAX16933 use a peak current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor, so the controller uses the voltage drop across the DC resistance of the inductor or the alternate series currentsense resistor to measure the inductor current. Currentmode control eliminates the double pole in the feedback loop caused by the inductor and output capacitor resulting in a smaller phase shift and requiring less elaborate error-amplifier compensation than voltage-mode control. A single series resistor (RC) and capacitor (CC) is all that is required to have a stable, high-bandwidth loop in applications where ceramic capacitors are used for output filtering (see Figure 2). For other types of capacitors, due to the higher capacitance and ESR, the frequency of the zero created by the capacitance and ESR is lower than the desired closed-loop crossover frequency. To stabilize a non-ceramic output capacitor loop, add another compensation capacitor (CF) from COMP to AGND to cancel this ESR zero. The basic regulator loop is modeled as a power modulator, output feedback divider, and an error amplifier as shown in Figure 2. The power modulator has a DC gain set by gmc x RLOAD, with a pole and zero pair set by RLOAD, the output capacitor (COUT), and its ESR. The loop response is set by the following equations: Figure 2. Compensation Network where RLOAD = VOUT/ILOUT(MAX) in and gmc =1/(AV_CS x RDC) in S. AV_CS is the voltage gain of the current-sense amplifier and is typically 11V/V. RDC is the DC resistance of the inductor or the current-sense resistor in . In a current-mode step-down converter, the output capacitor and the load resistance introduce a pole at the following frequency: f pMOD = 1 2 x C OUT x R LOAD The unity gain frequency of the power stage is set by COUT and gmc: f UGAINpMOD = g mc 2 x C OUT The output capacitor and its ESR also introduce a zero at: f zMOD = 1 2 x ESR x C OUT When COUT is composed of "n" identical capacitors in parallel, the resulting COUT = nxCOUT(EACH), and ESR = ESR(EACH) /n. Note that the capacitor zero for a parallel combination of alike capacitors is the same as for an individual capacitor. The feedback voltage-divider has a gain of GAINFB = VFB /VOUT, where VFB is 1V (typ). GAINMOD(dc) = g mc x R LOAD www.maximintegrated.com Maxim Integrated 18 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current The transconductance error amplifier has a DC gain of GAINEA(DC) = gm,EA x ROUT,EA, where gm,EA is the error amplifier transconductance, which is 1200S (typ), and ROUT,EA is the output resistance of the error amplifier, which is 30M (typ) (see the Electrical Characteristics table.) Set the error-amplifier compensation zero formed by RC and CC at the fpMOD. Calculate the value of CC as follows: A dominant pole (fdpEA) is set by the compensation capacitor (CC) and the amplifier output resistance (ROUT,EA). A zero (fZEA) is set by the compensation resistor (RC) and the compensation capacitor (CC). There is an optional pole (fPEA) set by CF and RC to cancel the output capacitor ESR zero if it occurs near the crossover frequency (fC, where the loop gain equals 1 (0dB)). Thus: If fzMOD is less than 5 x fC, add a second capacitor CF from COMP to AGND. The value of CF is: f dpEA = 1 2 x C C x (R OUT,EA + R C ) 1 f zEA = 2 x C C x R C f pEA = CF = As the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly and the crossover frequency remains the same. Below is a numerical example to calculate the compensation network component values of Figure 2: VOUT = 5V IOUT(MAX) = 5.33A RLOAD = VOUT /IOUT(MAX) = 5V/5.33A = 0.9375 COUT = 2x47F = 94F VFB x GAINEA(f ) = 1 C VOUT fSW = 26.4/65.5k = 0.403MHz GAINMOD(dc) = 6.06 x 0.9375 = 5.68 = f pMOD GAINEA(f = g m,EA x R C C) fC V GAINMOD(f ) x FB x g m,EA x R C = 1 C VOUT VOUT g m,EA x VFB x GAINMOD(f ) C 1 1.8kHz 2 x 94F x 0.9375 f f pMOD << f C SW 5 f pMOD Therefore: RC = 2 x f zMOD x R C ESR = 9m/2 = 4.5m At the crossover frequency, the total loop gain must be equal to 1. So: Solving for RC: 1 gmc = 1/(AV_CS x RDC) = 1/(11 x 0.015) = 6.06 f f pMOD << f C SW 5 C 2 x f pMOD x R C RDCR = 15m 1 2 x C F x R C GAIN = MOD(f ) GAINMOD(dc) x 1 AV_CS = 11V/V The loop-gain crossover frequency (fC) should be set below 1/5th of the switching frequency and much higher than the power-modulator pole (fpMOD). Select a value for fC in the range: GAINMOD(f ) x C CC = 1.8kHz << f C 80.6kHz select fC = 40kHz = f zMOD 1 376kHz 2 x 4.5m x 94F since fzMOD > fC: RC 16k CC 5.6nF CF 27pF www.maximintegrated.com Maxim Integrated 19 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Applications Information Layout Recommendations Careful PCB layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention (Figure 3). If possible, mount all the power components on the top side of the board, with their ground terminals flush against one another. Follow these guidelines for good PCB layout: * Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. * Keep the power traces and load connections short. This practice is essential for high efficiency. Using thick copper PCBs (2oz vs. 1oz) can enhance full-load efficiency by 1% or more. * Minimize current-sensing errors by connecting CS_ and OUT_. Use kelvin sensing directly across the current-sense resistor (RSENSE_). * Route high-speed switching nodes (BST_, LX_, DH_, and DL_) away from sensitive analog areas (FB_, CS_, and OUT_). Layout Procedure 1) Place the power components first, with ground terminals adjacent (low-side FET, CIN, COUT_, and Schottky). If possible, make all these connections on the top layer with wide, copper-filled areas. 2) Mount the controller IC adjacent to the low-side MOSFET, preferably on the back side opposite NL_ and NH_ to keep LX_, GND, DH_, and the DL_ gate drive lines short and wide. The DL_ and DH_ gate traces must be short and wide (50 mils to 100 mils wide if the MOSFET is 1in from the controller IC) to keep the driver impedance low and for proper adaptive dead-time sensing. 3) Group the gate-drive components (BST_ diode and capacitor and LDO bypass capacitor BIAS) together near the controller IC. Be aware that gate currents of up to 1A flow from the bootstrap capacitor to BST_, from DH_ to the gate of the external HS switch and from the LX_ pin to the inductor. Up to 100mA of current flow from the BIAS capacitor through the bootstrap diode to the bootstrap capacitor. Dimension those traces accordingly. 4) Make the DC-DC controller ground connections as shown in Figure 3. This diagram can be viewed as having two separate ground planes: power ground, where all the high-power components go; and an analog ground plane for sensitive analog components. The analog ground plane and power ground plane must meet only at a single point directly under the IC. 5) Connect the output power planes directly to the output filter capacitor positive and negative terminals with multiple vias. Place the entire DC-DC converter circuit as close to the load as is practical. KELVIN-SENSE VIAS UNDER THE SENSE RESISTOR (REFER TO THE EVALUATION KIT) INDUCTOR LOW-SIDE n-CHANNEL MOSFET (NH) COUT CIN COUT HIGH-SIDE n-CHANNEL MOSFET (NL) INPUT OUTPUT GROUND Figure 3. Layout Example www.maximintegrated.com Maxim Integrated 20 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Block Diagram PGOOD1 COMP1 DC-DC1 CONTROL LOGIC PGOOD LOW LEVEL PGOOD HIGH LEVEL PGOOD COMP FB1 FEEDBACK SELECT LOGIC MAX16932 MAX16933 EAMP1 INTERNAL SOFT-START EN1 REF = 1V OUT1 80 mV(TYP) MAX DIFFERENTIAL INPUT PWM1 PWM1 CSA1 CLK1 CS1 ZX1 CL SLOPE COMP LOGIC EN1 DH1 STEP-DOWN DC-DC1 GATE DRIVE LOGIC LX1 DL1 ZERO CROSS COMP CURRENT-LIMIT THRESHOLD LX1 FOSC BST1 PGND1 LX1 CLK1 OSCILLATOR IN SPREAD SPECTRUM OPTION AVAILABLE WITH INTERNAL CLOCK ONLY FSYNC FSYNC SELECT LOGIC EXTERNAL CLOCK INPUT BIAS INTERNAL LINEAR REGULATOR TIED HIGH (PWM MODE) TIED LOW (SKIP MODE) SWITCHOVER AGND EN2 COMP2 EN2 BST2 PWM2 FB2 CS2 IF 3.1V < VEXTVCC < 5.2V EXTVCC CLK 180 OUT-OF-PHASE CLK2 OUT2 BIAS CLK2 DC-DC2 CONTROL LOGIC SAME AS DC-DC1 ABOVE ZX2 PGOOD2 LX2 LX2 DH2 STEP-DOWN DC-DC2 GATE DRIVE LOGIC LX2 DL2 PGND2 EP www.maximintegrated.com Maxim Integrated 21 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Typical Operating Circuit RPGOOD1 FB1 OUT1 OUT1 CS1 CS1 PGOOD1 DL1 COMP1 AGND COMP2 OUT1 DH1 MAX16933 BIAS PGND1 EXTVCC IN OUT1 VBAT CIN DH2 EN2 BST2 CS2 CS2 LX2 OUT2 OUT2 FB2 COUT1 L1 BST1 PGND2 MAX16932 FOSC RCS1* LX1 EN1 FSYNC CS1 DL2 PGOOD2 RCS2* L2 CS2 OUT2 COUT2 PGND2 *DCR SENSE IS ALSO AN OPTION. www.maximintegrated.com Maxim Integrated 22 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Selector Guide BUCK 1 SWITCHING FREQUENCY (fSW1) BUCK 2 SWITCHING FREQUENCY (fSW2) SPREAD SPECTRUM (%) MAX16932ATIR/V+ 1MHz to 2.2MHz fSW1 -- MAX16932ATIT/V+ 1MHz to 2.2MHz 1/2fSW1 -- MAX16932CATIR/VY+ 1MHz to 2.2MHz fSW1 -- MAX16932CATIS/V+ 1MHz to 2.2MHz fSW1 6 MAX16932CATIS/VY+ 1MHz to 2.2MHz fSW1 6 MAX16932CATIU/V+ 1MHz to 2.2MHz 1/2fSW1 6 MAX16933ATIR/V+ 200kHz to 1MHz fSW1 -- MAX16933CATIR/VY+ 200kHz to 1MHz fSW1 -- MAX16933CATIS/V+ 200kHz to 1MHz fSW1 6 MAX16933CATIS/VY+ 200kHz to 1MHz fSW1 6 PART Chip Information Package Information For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a "+", "#", or "-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE PKG CODE OUTLINE NO. 28 TQFN-EP T2855+5 21-0140 LAND PATTERN NO. 90-0025 28 (SW) TQFN-EP T2855Y+6 21-1000041 90-0026 PROCESS: BiCMOS Ordering Information PART TEMP RANGE PIN-PACKAGE MAX16932ATI_/V+ -40C to +125C 28 TQFN-EP* MAX16932CATI_/V+ -40C to +125C 28 TQFN-EP* MAX16932CATI_/VY+ -40C to +125C 28 (SW) TQFN-EP* MAX16933ATI_/V+ -40C to +125C 28 TQFN-EP* MAX16933CATI_/V+ -40C to +125C 28 TQFN-EP* MAX16933CATI_/VY+ -40C to +125C 28 (SW) TQFN-EP* Note: Insert the desired suffix letter (from Selector Guide) into the blank to indicate buck 2 switching frequency and spread spectrum. /V denotes an automotive qualified part. +Denotes a lead(Pb)-free/RoHS-compliant package. (SW) = Side-wettable TQFN package. *EP = Exposed pad. www.maximintegrated.com Maxim Integrated 23 MAX16932/MAX16933 2.2MHz, 36V, Dual Buck with 20A Quiescent Current Revision History REVISION NUMBER REVISION DATE 0 6/13 Initial release -- 1 8/13 Duty Cycle Range in Electrical Characteristics changed to Maximum Duty Cycle and min/typ/max values updated 3 2 2/14 Updated PGOOD1 latchup references 10, 13, 22 3 1/15 Updated Benefits and Features section 1 4 7/15 Changed MAX16932ATIS/V+ to MAX16932BATIS/V+, MAX16932ATIU/V+ to MAX16932BATIU/V+, and MAX16933ATIS/V+ to MAX16933BATIS/V+, updating Electrical Characteristics, TOCs 25-27, Spread Spectrum and Buck 2 Switching Frequency sections, and the Selector Guide; added MAX16932BATI_/V+ and MAX16933BATI_/V+ future product variants to Ordering Information 5 8/15 Added spread-spectrum percentage for the MAX16932BATIS/V+ in Selector Guide 23 6 1/16 Removed future product designations in Ordering Information 23 7 4/16 Added future products (MAX16932CATIS/V+, MAX16932CATIU/V+, and MAX16933CATIS/V+) to the Selector Guide 23 6/16 Globally changed MAX16932B to MAX16932C and MAX16933B to MAX16933C in the Electrical Characteristics, Typical Operating Characteristics, Spread Spectrum, and Buck 2 Switching Frequency sections; removed the MAX16932BATIS/V+. MAX16932BATIU/V+, and MAX16933BATIS/V+ variants and deleted all future product asterisks in the Selector Guide; changed MAX16932BATI_V+ to MAX16932CATI_V+ and MAX16933BATI_V+ to MAX16933CATI_V+ in Ordering Information, and deleted the future parts footnote at the bottom of the table 9 6/17 Added MAX16932CATIR/VY+, MAX16932CATIS/VY+, MAX16933CATIR/VY+, and MAX16933CATIS/VY+ to the Selector Guide as future products; added MAX16932CATI_/VY+ and MAX16933CATI_/VY+ to Ordering Information, and 28 (SW) TQFN-EP to Package Information 23 10 9/17 Removed future product asterisks on the MAX16932CATIR/VY+, MAX16932CATIS/VY+, MAX16933CATIR/VY+, and MAX16933CATIS/VY+ in the Selector Guide 23 8 PAGES CHANGED DESCRIPTION 3, 8, 12, 23 3, 9, 12, 23 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated's website at www.maximintegrated.com. Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance. Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc. (c) 2017 Maxim Integrated Products, Inc. 24 Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Maxim Integrated: MAX16932ATIU/V+T MAX16932ATIU/V+ MAX16932ATIR/V+ MAX16932ATIT/V+ MAX16932ATIS/V+ MAX16932ATIT/V+T MAX16932ATIR/V+T MAX16933ATIS/V+T MAX16932ATIS/V+T MAX16933ATIR/V+ MAX16933ATIR/V+T