LM3410XBSTOVPEV/NOPB - NATIONAL SEMICONDUCTOR

LM3410

LM3410/LM3410Q 525kHz/1.6MHz, Constant Current Boost and SEPIC LED Driver

with Internal Compensation

Literature Number: SNVS541F

LM3410/LM3410Q

May 18, 2009

525kHz/1.6MHz, Constant Current Boost and SEPIC LED

Driver with Internal Compensation

General Description

The LM3410 constant current LED driver is a monolithic, high

frequency, PWM DC/DC converter in 5-pin SOT23, 6-pin LLP,

& 8-pin eMSOP packages. With a minimum of external com-

ponents the LM3410 is easy to use. It can drive 2.8A typical

peak currents with an internal 170 mΩ NMOS switch. Switch-

ing frequency is internally set to either 525 kHz or 1.60 MHz,

allowing the use of extremely small surface mount inductors

and chip capacitors. Even though the operating frequency is

high, efficiencies up to 88% are easy to achieve. External

shutdown is included, featuring an ultra-low standby current

of 80 nA. The LM3410 utilizes current-mode control and in-

ternal compensation to provide high-performance over a wide

range of operating conditions. Additional features include

dimming, cycle-by-cycle current limit, and thermal shutdown.

Features

■Space Saving SOT23-5 & 6-LLP Package

■Input voltage range of 2.7V to 5.5V

■Output voltage range of 3V to 24V

■2.8A Typical Switch Current

■High Switching Frequency

—525 KHz (LM3410-Y)

—1.6 MHz (LM3410-X)

■170 mΩ NMOS Switch

■190 mV Internal Voltage Reference

■Internal Soft-Start

■Current-Mode, PWM Operation

■Thermal Shutdown

■LM3410Q is AEC-Q100 Grade 1 qualified and is

manufactured on an Automotive Grade Flow

Applications

■LED Backlight Current Source

■LiIon Backlight OLED & HB LED Driver

■Handheld Devices

■LED Flash Driver

■Automotive

Typical Boost Application Circuit

30038501

30038502

LM3410/LM3410Q 525kHz/1.6MHz, Constant Current Boost and SEPIC LED Driver with Internal

Compensation

Connection Diagrams

Top View

30038503

5–Pin SOT23

Top View

30038504

6-Pin LLP

Top View

30038505

8-Pin eMSOP

Ordering Information

Order Number Frequency Package Type Package Drawing Supplied As Feature

LM3410YMF

525 kHz

SOT23-5 MF05A

1000 units Tape & Reel

LM3410YMFX 3000 units Tape & Reel

LM3410YMFE 250 units Tape & Reel

LM3410YQMF 1000 units Tape & Reel AEC-Q100 Grade 1

qualified. Automotive

Grade Production

Flow*

LM3410YQMFX 3000 units Tape & Reel

LM3410YSD

LLP-6 SDE06A

1000 units Tape & Reel

LM3410YSDX 4500 units Tape & Reel

LM3410YSDE 250 units Tape & Reel

LM3410YMY

eMSOP-8 MUY08A

1000 units Tape & Reel

LM3410YMYX 3500 units Tape & Reel

LM3410YMYE 250 units Tape & Reel

LM3410XMF

1.6 MHz

SOT23-5 MF05A

1000 units Tape & Reel

LM3410XMFX 3000 units Tape & Reel

LM3410XMFE 250 units Tape & Reel

LM3410XQMF 1000 units Tape & Reel AEC-Q100 Grade 1

qualified. Automotive

Grade Production

Flow*

LM3410XQMFX 3000 units Tape & Reel

LM3410XSD

LLP-6 SDE06A

1000 units tape & reel

LM3410XSDX 4500 units Tape & Reel

LM3410XSDE 250 units Tape & Reel

LM3410XMY

eMSOP-8 MUY08A

1000 units Tape & Reel

LM3410XMYX 3500 units Tape & Reel

LM3410XMYE 250 units Tape & Reel

*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.

Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified

with the letter Q. For more information go to http://www.national.com/automotive.

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LM3410/LM3410Q

Pin Descriptions - 5-Pin SOT23

Pin Name Function

1 SW Output switch. Connect to the inductor, output diode.

2 GND Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this

pin.

3 FB Feedback pin. Connect FB to external resistor divider to set output voltage.

4 DIM Dimming & shutdown control input. Logic high enables operation. Duty Cycle from 0 to 100%. Do not allow

this pin to float or be greater than VIN + 0.3V.

5 VIN Supply voltage pin for power stage, and input supply voltage.

Pin Descriptions - 6-Pin LLP

Pin Name Function

1 PGND Power ground pin. Place PGND and output capacitor GND close together.

2 VIN Supply voltage for power stage, and input supply voltage.

3 DIM Dimming & shutdown control input. Logic high enables operation. Duty Cycle from 0 to 100%. Do not allow

this pin to float or be greater than VIN + 0.3V.

4 FB Feedback pin. Connect FB to external resistor divider to set output voltage.

5 AGND Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin

6 SW Output switch. Connect to the inductor, output diode.

DAP GND Signal & Power ground. Connect to pin 1 & pin 5 on top layer. Place 4-6 vias from DAP to bottom layer GND

plane.

Pin Descriptions - 8-Pin eMSOP

Pin Name Function

1 - No Connect

2 PGND Power ground pin. Place PGND and output capacitor GND close together.

3 VIN Supply voltage for power stage, and input supply voltage.

4 DIM Dimming & shutdown control input. Logic high enables operation. Duty Cycle from 0 to 100%. Do not allow

this pin to float or be greater than VIN + 0.3V.

5 FB Feedback pin. Connect FB to external resistor divider to set output voltage.

6 AGND Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin 5

7 SW Output switch. Connect to the inductor, output diode.

8 - No Connect

DAP GND Signal & Power ground. Connect to pin 2 & pin 6 on top layer. Place 4-6 vias from DAP to bottom layer GND

plane.

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LM3410/LM3410Q

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN -0.5V to 7V

SW Voltage -0.5V to 26.5V

FB Voltage -0.5V to 3.0V

DIM Voltage -0.5V to 7.0V

ESD Susceptibility (Note 4)

Human Body Model 2kV

Junction Temperature (Note 2) 150°C

Storage Temp. Range -65°C to 150°C

Soldering Information

Infrared/Convection Reflow (15sec) 220°C

Operating Ratings (Note 1)

VIN 2.7V to 5.5V

VDIM (Note 5) 0V to VIN

VSW 3V to 24V

Junction Temperature Range -40°C to 125°C

Power Dissipation

(Internal) SOT23-5 400 mW

Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the

junction temperature (TJ) range of -40°C to 125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical

correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.

VIN = 5V, unless otherwise indicated under the Conditions column.

Symbol Parameter Conditions Min Typ Max Units

VFB Feedback Voltage 178 190 202 mV

ΔVFB/VIN Feedback Voltage Line Regulation VIN = 2.7V to 5.5V - 0.06 - %/V

IFB Feedback Input Bias Current - 0.1 1µA

FSW Switching Frequency LM3410-X 1200 1600 2000 kHz

LM3410-Y 360 525 680

DMAX Maximum Duty Cycle LM3410-X 88 92 - %

LM3410-Y 90 95 -

DMIN Minimum Duty Cycle LM3410-X - 5 - %

LM3410-Y - 2 -

RDS(ON) Switch On Resistance SOT23-5 and eMSOP-8 - 170 330 mΩ

LLP-6 190 350

ICL Switch Current Limit 2.1 2.80 - A

SU Start Up Time - 20 - µs

Quiescent Current (switching) LM3410-X VFB = 0.25 - 7.0 11 mA

LM3410-Y VFB = 0.25 - 3.4 7

Quiescent Current (shutdown) All Options VDIM = 0V - 80 - nA

UVLO Undervoltage Lockout VIN Rising - 2.3 2.65 V

VIN Falling 1.7 1.9 -

VDIM_H

Shutdown Threshold Voltage - - 0.4 V

Enable Threshold Voltage 1.8 - -

ISW Switch Leakage VSW = 24V - 1.0 - µA

IDIM Dimming Pin Current Sink/Source - 100 - nA

θJA

Junction to Ambient

0 LFPM Air Flow (Note 3)

LLP-6 and eMSOP-8 Package - 80 - °C/W

SOT23-5 Package - 118 -

θJC Junction to Case (Note 3)LLP-6 and eMSOP-8 Package - 18 - °C/W

SOT23-5 Package - 60 -

TSD Thermal Shutdown Temperature (Note 2) - 165 - °C

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but does not guarantee specific performance limits. For guaranteed specifications and conditions, see the Electrical Characteristics.

Note 2: Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device.

Note 3: Applies for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air.

Note 4: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test method is per JESD22-A114.

Note 5: Do not allow this pin to float or be greater than VIN +0.3V.

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LM3410/LM3410Q

Typical Performance Characteristics All curves taken at VIN = 5.0V with configuration in typical

application circuit shown in Application Information section of this datasheet. TJ = 25C, unless otherwise specified.

LM3410-X Efficiency vs VIN (RSET = 4Ω)

30038502

LM3410-X Start-Up Signature

30038507

4 x 3.3V LEDs 500 Hz DIM FREQ D = 50%

30038508

DIM Freq & Duty Cycle vs Avg I-LED

30038509

Current Limit vs Temperature

30038510

RDSON vs Temperature

30038511

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LM3410/LM3410Q

Oscillator Frequency vs Temperature - "X"

30038512

Oscillator Frequency vs Temperature - "Y"

30038513

VFB vs Temperature

30038580

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LM3410/LM3410Q

Simplified Internal Block Diagram

30038514

FIGURE 1. Simplified Block Diagram

Application Information

THEORY OF OPERATION

The LM3410 is a constant frequency PWM, boost regulator

IC. It delivers a minimum of 2.1A peak switch current. The

device operates very similar to a voltage regulated boost con-

verter except that it regulates the output current through

LEDs. The current magnitude is set with a series resistor. This

series resistor multiplied by the LED current creates the feed-

back voltage (190 mV) which the converter regulates to. The

regulator has a preset switching frequency of either 525 kHz

or 1.60 MHz. This high frequency allows the LM3410 to op-

erate with small surface mount capacitors and inductors,

resulting in a DC/DC converter that requires a minimum

amount of board space. The LM3410 is internally compen-

sated, so it is simple to use, and requires few external com-

ponents. The LM3410 uses current-mode control to regulate

the LED current. The following operating description of the

LM3410 will refer to the Simplified Block Diagram (Figure 1)

the simplified schematic (Figure 2), and its associated wave-

forms (Figure 3). The LM3410 supplies a regulated LED

current by switching the internal NMOS control switch at con-

stant frequency and variable duty cycle. A switching cycle

begins at the falling edge of the reset pulse generated by the

internal oscillator. When this pulse goes low, the output con-

trol logic turns on the internal NMOS control switch. During

this on-time, the SW pin voltage (VSW) decreases to approx-

imately GND, and the inductor current (IL) increases with a

linear slope. IL is measured by the current sense amplifier,

which generates an output proportional to the switch current.

The sensed signal is summed with the regulator’s corrective

ramp and compared to the error amplifier’s output, which is

proportional to the difference between the feedback voltage

and VREF. When the PWM comparator output goes high, the

output switch turns off until the next switching cycle begins.

During the switch off-time, inductor current discharges

through diode D1, which forces the SW pin to swing to the

output voltage plus the forward voltage (VD) of the diode. The

regulator loop adjusts the duty cycle (D) to maintain a regu-

lated LED current.

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LM3410/LM3410Q

30038515

FIGURE 2. Simplified Boost Topology Schematic

30038516

FIGURE 3. Typical Waveforms

CURRENT LIMIT

The LM3410 uses cycle-by-cycle current limiting to protect

the internal NMOS switch. It is important to note that this cur-

rent limit will not protect the output from excessive current

during an output short circuit. The input supply is connected

to the output by the series connection of an inductor and a

diode. If a short circuit is placed on the output, excessive cur-

rent can damage both the inductor and diode.

Design Guide

SETTING THE LED CURRENT

30038517

FIGURE 4. Setting ILED

The LED current is set using the following equation:

where RSET is connected between the FB pin and GND.

DIM PIN / SHUTDOWN MODE

The average LED current can be controlled using a PWM

signal on the DIM pin. The duty cycle can be varied between

0 & 100% to either increase or decrease LED brightness.

PWM frequencies in the range of 1 Hz to 25 kHz can be used.

For controlling LED currents down to the µA levels, it is best

to use a PWM signal frequency between 200-1 kHz. The

maximum LED current would be achieved using a 100% duty

cycle, i.e. the DIM pin always high.

LED-DRIVE CAPABILITY

When using the LM3410 in the typical application configura-

tion, with LEDs stacked in series between the VOUT and FB

pin, the maximum number of LEDs that can be placed in se-

ries is dependent on the maximum LED forward voltage

(VFMAX).

(VFMAX x NLEDs) + 190 mV < 24V

When inserting a value for maximum VFMAX the LED forward

voltage variation over the operating temperature range

should be considered.

THERMAL SHUTDOWN

Thermal shutdown limits total power dissipation by turning off

the output switch when the IC junction temperature exceeds

165°C. After thermal shutdown occurs, the output switch

doesn’t turn on until the junction temperature drops to ap-

proximately 150°C.

INDUCTOR SELECTION

The inductor value determines the input ripple current. Lower

inductor values decrease the physical size of the inductor, but

increase the input ripple current. An increase in the inductor

value will decrease the input ripple current.

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LM3410/LM3410Q

30038519

FIGURE 5. Inductor Current

The Duty Cycle (D) for a Boost converter can be approximat-

ed by using the ratio of output voltage (VOUT) to input voltage

(VIN).

Therefore:

Power losses due to the diode (D1) forward voltage drop, the

voltage drop across the internal NMOS switch, the voltage

drop across the inductor resistance (RDCR) and switching

losses must be included to calculate a more accurate duty

cycle (See Calculating Efficiency and Junction Tempera-

ture for a detailed explanation). A more accurate formula for

calculating the conversion ratio is:

Where η equals the efficiency of the LM3410 application.

Or:

Therefore:

Inductor ripple in a LED driver circuit can be greater than what

would normally be allowed in a voltage regulator Boost &

Sepic design. A good design practice is to allow the inductor

to produce 20% to 50% ripple of maximum load. The in-

creased ripple shouldn’t be a problem when illuminating

LEDs.

From the previous equations, the inductor value is then ob-

tained.

Where

1/TS = fSW

One must also ensure that the minimum current limit (2.1A)

is not exceeded, so the peak current in the inductor must be

calculated. The peak current (Lpk I) in the inductor is calcu-

lated by:

ILpk = IIN + ΔIL or ILpk = IOUT /D' + ΔiL

When selecting an inductor, make sure that it is capable of

supporting the peak input current without saturating. Inductor

saturation will result in a sudden reduction in inductance and

prevent the regulator from operating correctly. Because of the

speed of the internal current limit, the peak current of the in-

ductor need only be specified for the required maximum input

current. For example, if the designed maximum input current

is 1.5A and the peak current is 1.75A, then the inductor should

be specified with a saturation current limit of >1.75A. There is

no need to specify the saturation or peak current of the in-

ductor at the 2.8A typical switch current limit.

Because of the operating frequency of the LM3410, ferrite

based inductors are preferred to minimize core losses. This

presents little restriction since the variety of ferrite-based in-

ductors is huge. Lastly, inductors with lower series resistance

(DCR) will provide better operating efficiency. For recom-

mended inductors see Example Circuits.

INPUT CAPACITOR

An input capacitor is necessary to ensure that VIN does not

drop excessively during switching transients. The primary

specifications of the input capacitor are capacitance, voltage,

RMS current rating, and ESL (Equivalent Series Inductance).

The recommended input capacitance is 2.2 µF to 22 µF de-

pending on the application. The capacitor manufacturer

specifically states the input voltage rating. Make sure to check

any recommended deratings and also verify if there is any

significant change in capacitance at the operating input volt-

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LM3410/LM3410Q

age and the operating temperature. The ESL of an input

capacitor is usually determined by the effective cross sec-

tional area of the current path. At the operating frequencies

of the LM3410, certain capacitors may have an ESL so large

that the resulting impedance (2πfL) will be higher than that

required to provide stable operation. As a result, surface

mount capacitors are strongly recommended. Multilayer ce-

ramic capacitors (MLCC) are good choices for both input and

output capacitors and have very low ESL. For MLCCs it is

recommended to use X7R or X5R dielectrics. Consult capac-

itor manufacturer datasheet to see how rated capacitance

varies over operating conditions.

OUTPUT CAPACITOR

The LM3410 operates at frequencies allowing the use of ce-

ramic output capacitors without compromising transient re-

sponse. Ceramic capacitors allow higher inductor ripple

without significantly increasing output ripple. The output ca-

pacitor is selected based upon the desired output ripple and

transient response. The initial current of a load transient is

provided mainly by the output capacitor. The output

impedance will therefore determine the maximum voltage

perturbation. The output ripple of the converter is a function

of the capacitor’s reactance and its equivalent series resis-

tance (ESR):

When using MLCCs, the ESR is typically so low that the ca-

pacitive ripple may dominate. When this occurs, the output

ripple will be approximately sinusoidal and 90° phase shifted

from the switching action.

Given the availability and quality of MLCCs and the expected

output voltage of designs using the LM3410, there is really no

need to review any other capacitor technologies. Another

benefit of ceramic capacitors is their ability to bypass high

frequency noise. A certain amount of switching edge noise

will couple through parasitic capacitances in the inductor to

the output. A ceramic capacitor will bypass this noise while a

tantalum will not. Since the output capacitor is one of the two

external components that control the stability of the regulator

control loop, most applications will require a minimum at 0.47

µF of output capacitance. Like the input capacitor, recom-

mended multilayer ceramic capacitors are X7R or X5R.

Again, verify actual capacitance at the desired operating volt-

age and temperature.

DIODE

The diode (D1) conducts during the switch off time. A Schottky

diode is recommended for its fast switching times and low

forward voltage drop. The diode should be chosen so that its

current rating is greater than:

ID1 ≥ IOUT

The reverse breakdown rating of the diode must be at least

the maximum output voltage plus appropriate margin.

OUTPUT OVER-VOLTAGE PROTECTION

A simple circuit consisting of an external zener diode can be

implemented to protect the output and the LM3410 device

from an over-voltage fault condition. If an LED fails open, or

is connected backwards, an output open circuit condition will

occur. No current is conducted through the LED’s, and the

feedback node will equal zero volts. The LM3410 will react to

this fault by increasing the duty-cycle, thinking the LED cur-

rent has dropped. A simple circuit that protects the LM3410

is shown in figure 6.

Zener diode D2 and resistor R3 is placed from VOUT in parallel

with the string of LEDs. If the output voltage exceeds the

breakdown voltage of the zener diode, current is drawn

through the zener diode, R3 and sense resistor R1. Once the

voltage across R1 and R3 equals the feedback voltage of

190mV, the LM3410 will limit its duty-cycle. No damage will

occur to the LM3410, the LED’s, or the zener diode. Once the

fault is corrected, the application will work as intended.

30038530

FIGURE 6. Overvoltage Protection Circuit

PCB Layout Considerations

When planning layout there are a few things to consider when

trying to achieve a clean, regulated output. The most impor-

tant consideration when completing a Boost Converter layout

is the close coupling of the GND connections of the COUT ca-

pacitor and the LM3410 PGND pin. The GND ends should be

close to one another and be connected to the GND plane with

at least two through-holes. There should be a continuous

ground plane on the bottom layer of a two-layer board except

under the switching node island. The FB pin is a high

impedance node and care should be taken to make the FB

trace short to avoid noise pickup and inaccurate regulation.

The RSET feedback resistor should be placed as close as

possible to the IC, with the AGND of RSET (R1) placed as close

as possible to the AGND (pin 5 for the LLP) of the IC. Radiated

noise can be decreased by choosing a shielded inductor. The

remaining components should also be placed as close as

possible to the IC. Please see Application Note AN-1229 for

further considerations and the LM3410 demo board as an ex-

ample of a four-layer layout.

Below is an example of a good thermal & electrical PCB de-

sign.

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LM3410/LM3410Q

30038532

FIGURE 7. Boost PCB Layout Guidelines

This is very similar to our LM3410 demonstration boards that

are obtainable via the National Semiconductor website. The

demonstration board consists of a two layer PCB with a com-

mon input and output voltage application. Most of the routing

is on the top layer, with the bottom layer consisting of a large

ground plane. The placement of the external components

satisfies the electrical considerations, and the thermal perfor-

mance has been improved by adding thermal vias and a top

layer “Dog-Bone”.

For certain high power applications, the PCB land may be

modified to a "dog bone" shape (see Figure 8). Increasing the

size of ground plane and adding thermal vias can reduce the

RθJA for the application.

30038533

FIGURE 8. PCB Dog Bone Layout

Thermal Design

When designing for thermal performance, one must consider

many variables:

Ambient Temperature: The surrounding maximum air tem-

perature is fairly explanatory. As the temperature increases,

the junction temperature will increase. This may not be linear

though. As the surrounding air temperature increases, resis-

tances of semiconductors, wires and traces increase. This will

decrease the efficiency of the application, and more power

will be converted into heat, and will increase the silicon junc-

tion temperatures further.

Forced Airflow: Forced air can drastically reduce the device

junction temperature. Air flow reduces the hot spots within a

design. Warm airflow is often much better than a lower am-

bient temperature with no airflow.

External Components: Choose components that are effi-

cient, and you can reduce the mutual heating between de-

vices.

PCB design with thermal performance in mind:

The PCB design is a very important step in the thermal design

procedure. The LM3410 is available in three package options

(5 pin SOT23, 8 pin eMSOP & 6 pin LLP). The options are

electrically the same, but difference between the packages is

size and thermal performance. The LLP and eMSOP have

thermal Die Attach Pads (DAP) attached to the bottom of the

packages, and are therefore capable of dissipating more heat

than the SOT23 package. It is important that the customer

choose the correct package for the application. A detailed

thermal design procedure has been included in this data

sheet. This procedure will help determine which package is

correct, and common applications will be analyzed.

There is one significant thermal PCB layout design consider-

ation that contradicts a proper electrical PCB layout design

consideration. This contradiction is the placement of external

components that dissipate heat. The greatest external heat

contributor is the external Schottky diode. It would be nice if

you were able to separate by distance the LM3410 from the

Schottky diode, and thereby reducing the mutual heating ef-

fect. This will however create electrical performance issues.

It is important to keep the LM3410, the output capacitor, and

Schottky diode physically close to each other (see PCB layout

guidelines). The electrical design considerations outweigh the

thermal considerations. Other factors that influence thermal

performance are thermal vias, copper weight, and number of

board layers.

Thermal Definitions

Heat energy is transferred from regions of high temperature

to regions of low temperature via three basic mechanisms:

radiation, conduction and convection.

Radiation: Electromagnetic transfer of heat between masses

at different temperatures.

Conduction: Transfer of heat through a solid medium.

Convection: Transfer of heat through the medium of a fluid;

typically air.

Conduction & Convection will be the dominant heat transfer

mechanism in most applications.

RθJA: Thermal impedance from silicon junction to ambient air

temperature.

RθJC: Thermal impedance from silicon junction to device case

temperature.

CθJC: Thermal Delay from silicon junction to device case tem-

perature.

CθCA: Thermal Delay from device case to ambient air tem-

perature.

RθJA & RθJC: These two symbols represent thermal

impedances, and most data sheets contain associated values

for these two symbols. The units of measurement are °C/

Watt.

RθJA is the sum of smaller thermal impedances (see simplified

thermal model Figures 9 and 10). Capacitors within the model

represent delays that are present from the time that power

and its associated heat is increased or decreased from steady

state in one medium until the time that the heat increase or

decrease reaches steady state in the another medium.

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LM3410/LM3410Q

The datasheet values for these symbols are given so that one

might compare the thermal performance of one package

against another. To achieve a comparison between pack-

ages, all other variables must be held constant in the com-

parison (PCB size, copper weight, thermal vias, power

dissipation, VIN, VOUT, load current etc). This does shed light

on the package performance, but it would be a mistake to use

these values to calculate the actual junction temperature in

your application.

LM3410 Thermal Models

Heat is dissipated from the LM3410 and other devices. The

external loss elements include the Schottky diode, inductor,

and loads. All loss elements will mutually increase the heat

on the PCB, and therefore increase each other’s tempera-

tures.

30038534

FIGURE 9. Thermal Schematic

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LM3410/LM3410Q

30038535

FIGURE 10. Associated Thermal Model

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LM3410/LM3410Q

Calculating Efficiency, and Junction

Temperature

We will talk more about calculating proper junction tempera-

ture with relative certainty in a moment. For now we need to

describe how to calculate the junction temperature and clarify

some common misconceptions.

A common error when calculating RθJA is to assume that the

package is the only variable to consider.

RθJA [variables]:

•Input Voltage, Output Voltage, Output Current, RDS(ON)

•Ambient temperature & air flow

•Internal & External components power dissipation

•Package thermal limitations

•PCB variables (copper weight, thermal via’s, layers

component placement)

Another common error when calculating junction temperature

is to assume that the top case temperature is the proper tem-

perature when calculating RθJC. RθJC represents the thermal

impedance of all six sides of a package, not just the top side.

This document will refer to a thermal impedance called .

represents a thermal impedance associated with just the

top case temperature. This will allow one to calculate the

junction temperature with a thermal sensor connected to the

top case.

The complete LM3410 Boost converter efficiency can be cal-

culated in the following manner.

Power loss (PLOSS) is the sum of two types of losses in the

converter, switching and conduction. Conduction losses usu-

ally dominate at higher output loads, where as switching

losses remain relatively fixed and dominate at lower output

loads.

Losses in the LM3410 Device: PLOSS = PCOND + PSW + PQ

Where PQ = quiescent operating power loss

Conversion ratio of the Boost Converter with conduction loss

elements inserted:

Where

RDCR = Inductor series resistance

One can see that if the loss elements are reduced to zero, the

conversion ratio simplifies to:

And we know:

Therefore:

Calculations for determining the most significant power loss-

es are discussed below. Other losses totaling less than 2%

are not discussed.

A simple efficiency calculation that takes into account the

conduction losses is shown below:

The diode, NMOS switch, and inductor DCR losses are in-

cluded in this calculation. Setting any loss element to zero will

simplify the equation.

VD is the forward voltage drop across the Schottky diode. It

can be obtained from the manufacturer’s Electrical Charac-

teristics section of the data sheet.

The conduction losses in the diode are calculated as follows:

PDIODE = VD x ILED

Depending on the duty cycle, this can be the single most sig-

nificant power loss in the circuit. Care should be taken to

choose a diode that has a low forward voltage drop. Another

concern with diode selection is reverse leakage current. De-

pending on the ambient temperature and the reverse voltage

across the diode, the current being drawn from the output to

the NMOS switch during time D could be significant, this may

increase losses internal to the LM3410 and reduce the overall

efficiency of the application. Refer to Schottky diode

manufacturer’s data sheets for reverse leakage specifica-

tions, and typical applications within this data sheet for diode

selections.

Another significant external power loss is the conduction loss

in the input inductor. The power loss within the inductor can

be simplified to:

www.national.com 14

LM3410/LM3410Q

PIND = IIN2RDCR

The LM3410 conduction loss is mainly associated with the

internal power switch:

PCOND-NFET = I2SW-rms x RDSON x D

30038542

FIGURE 11. LM3410 Switch Current

(small ripple approximation)

PCOND-NFET = IIN2 x RDSON x D

The value for RDSON should be equal to the resistance at the

junction temperature you wish to analyze. As an example, at

125°C and RDSON = 250 mΩ (See typical graphs for value).

Switching losses are also associated with the internal power

switch. They occur during the switch on and off transition pe-

riods, where voltages and currents overlap resulting in power

loss.

The simplest means to determine this loss is to empirically

measuring the rise and fall times (10% to 90%) of the switch

at the switch node:

PSWR = 1/2(VOUT x IIN x fSW x tRISE)

PSWF = 1/2(VOUT x IIN x fSW x tFALL)

PSW = PSWR + PSWF

Typical Switch-Node Rise and Fall Times

VIN VOUT tRISE tFALL

3V 5V 6nS 4nS

5V 12V 6nS 5nS

3V 12V 8nS 7nS

5V 18V 10nS 8nS

Quiescent Power Losses

IQ is the quiescent operating current, and is typically around

1.5 mA.

PQ = IQ x VIN

RSET Power Loss

Example Efficiency Calculation:

Operating Conditions:

5 x 3.3V LEDs + 190mVREF ≊ 16.7V

TABLE 1. Operating Conditions

VIN 3.3V

VOUT 16.7V

ILED 50mA

VD0.45V

fSW 1.60MHz

IQ3mA

tRISE 10nS

tFALL 10nS

RDSON 225mΩ

LDCR 75mΩ

D0.82

IIN 0.31A

ΣPCOND + PSW + PDIODE + PIND + PQ = PLOSS

Quiescent Power Loss:

PQ = IQ x VIN = 10 mW

Switching Power Loss:

PSWR = 1/2(VOUT x IIN x fSW x tRISE) ≊ 40 mW

PSWF = 1/2(VOUT x IIN x fSW x tFALL) ≊ 40 mW

PSW = PSWR + PSWF = 80 mW

Internal NFET Power Loss:

RDSON = 225 mΩ

PCONDUCTION = IIN2 x D x RDSON = 17 mW

IIN = 310 mA

Diode Loss:

VD = 0.45V

PDIODE = VD x ILED = 23 mW

Inductor Power Loss:

RDCR = 75 mΩ

PIND = IIN2 x RDCR = 7 mW

15 www.national.com

LM3410/LM3410Q

Total Power Losses are:

TABLE 2. Power Loss Tabulation

VIN 3.3V

VOUT 16.7V

ILED 50mA POUT 825W

VD0.45V PDIODE 23mW

fSW 1.6MHz

IQ10nS PSWR 40mW

tRISE 10nS PSWF 40mW

IQ3mA PQ10mW

RDSON 225mΩPCOND 17mW

LDCR 75mΩPIND 7mW

D0.82

η85% PLOSS 137mW

PINTERNAL = PCOND + PSW = 107 mW

Calculating and

We now know the internal power dissipation, and we are try-

ing to keep the junction temperature at or below 125°C. The

next step is to calculate the value for and/or . This is

actually very simple to accomplish, and necessary if you think

you may be marginal with regards to thermals or determining

what package option is correct.

The LM3410 has a thermal shutdown comparator. When the

silicon reaches a temperature of 165°C, the device shuts

down until the temperature drops to 150°C. Knowing this, one

can calculate the or the of a specific application. Be-

cause the junction to top case thermal impedance is much

lower than the thermal impedance of junction to ambient air,

the error in calculating is lower than for . However,

you will need to attach a small thermocouple onto the top case

of the LM3410 to obtain the value.

Knowing the temperature of the silicon when the device shuts

down allows us to know three of the four variables. Once we

calculate the thermal impedance, we then can work back-

wards with the junction temperature set to 125°C to see what

maximum ambient air temperature keeps the silicon below

the 125°C temperature.

Procedure:

Place your application into a thermal chamber. You will need

to dissipate enough power in the device so you can obtain a

good thermal impedance value.

Raise the ambient air temperature until the device goes into

thermal shutdown. Record the temperatures of the ambient

air and/or the top case temperature of the LM3410. Calculate

the thermal impedances.

Example from previous calculations (SOT23-5 Package):

PINTERNAL = 107 mW

TA @ Shutdown = 155°C

TC @ Shutdown = 159°C

SOT23-5 = 93°C/W

SOT23-5 = 56°C/W

Typical LLP & eMSOP typical applications will produce

numbers in the range of 50°C/W to 65°C/W, and will vary

between 18°C/W and 28°C/W. These values are for PCB’s

with two and four layer boards with 0.5 oz copper, and four to

six thermal vias to bottom side ground plane under the DAP.

The thermal impedances calculated above are higher due to

the small amount of power being dissipated within the device.

Note: To use these procedures it is important to dissipate an

amount of power within the device that will indicate a true

thermal impedance value. If one uses a very small internal

dissipated value, one can see that the thermal impedance

calculated is abnormally high, and subject to error. Figure 12

shows the nonlinear relationship of internal power dissipation

vs . .

30038551

FIGURE 12. RθJA vs Internal Dissipation

For 5-pin SOT23 package typical applications, RθJA numbers

will range from 80°C/W to 110°C/W, and will vary between

50°C/W and 65°C/W. These values are for PCB’s with two &

four layer boards with 0.5 oz copper, with two to four thermal

vias from GND pin to bottom layer.

Here is a good rule of thumb for typical thermal impedances,

and an ambient temperature maximum of 75°C: If your design

requires that you dissipate more than 400mW internal to the

LM3410, or there is 750mW of total power loss in the appli-

cation, it is recommended that you use the 6 pin LLP or the 8

pin eMSOP package with the exposed DAP.

SEPIC Converter

The LM3410 can easily be converted into a SEPIC converter.

A SEPIC converter has the ability to regulate an output volt-

age that is either larger or smaller in magnitude than the input

voltage. Other converters have this ability as well (CUK and

Buck-Boost), but usually create an output voltage that is op-

posite in polarity to the input voltage. This topology is a perfect

fit for Lithium Ion battery applications where the input voltage

for a single cell Li-Ion battery will vary between 2.7V & 4.5V

and the output voltage is somewhere in between. Most of the

www.national.com 16

LM3410/LM3410Q

analysis of the LM3410 Boost Converter is applicable to the

LM3410 SEPIC Converter.

SEPIC Design Guide:

SEPIC Conversion ratio without loss elements:

Therefore:

Small ripple approximation:

In a well-designed SEPIC converter, the output voltage, and

input voltage ripple, the inductor ripple IL1 and IL2 is small in

comparison to the DC magnitude. Therefore it is a safe ap-

proximation to assume a DC value for these components. The

main objective of the Steady State Analysis is to determine

the steady state duty-cycle, voltage and current stresses on

all components, and proper values for all components.

In a steady-state converter, the net volt-seconds across an

inductor after one cycle will equal zero. Also, the charge into

a capacitor will equal the charge out of a capacitor in one cy-

cle.

Therefore:

Substituting IL1 into IL2

IL2 = ILED

The average inductor current of L2 is the average output load.

30038556

FIGURE 13. Inductor Volt-Sec Balance Waveform

Applying Charge balance on C1:

Since there are no DC voltages across either inductor, and

capacitor C3 is connected to Vin through L1 at one end, or to

ground through L2 on the other end, we can say that

VC3 = VIN

Therefore:

This verifies the original conversion ratio equation.

It is important to remember that the internal switch current is

equal to IL1 and IL2 during the D interval. Design the converter

so that the minimum guaranteed peak switch current limit

(2.1A) is not exceeded.

30038552

FIGURE 14. HB/OLED SEPIC CONVERTER Schematic

17 www.national.com

LM3410/LM3410Q

Steady State Analysis with Loss

Elements

30038559

FIGURE 15. SEPIC Simplified Schematic

Using inductor volt-second balance & capacitor charge bal-

ance, the following equations are derived:

IL2 = (ILED)

and

IL1 = (ILED) x (D/D')

Therefore:

One can see that all variables are known except for the duty

cycle (D). A quadratic equation is needed to solve for D. A

less accurate method of determining the duty cycle is to as-

sume efficiency, and calculate the duty cycle.

TABLE 3. Efficiencies for Typical SEPIC Applications

VIN 2.7V VIN 3.3V VIN 5V

VOUT 3.1V VOUT 3.1V VOUT 3.1V

IIN 770mA IIN 600mA IIN 375mA

ILED 500mA ILED 500mA ILED 500mA

η75% η80% η83%

SEPIC Converter PCB Layout

The layout guidelines described for the LM3410 Boost-Con-

verter are applicable to the SEPIC OLED Converter. Figure

16 is a proper PCB layout for a SEPIC Converter.

30038565

FIGURE 16. HB/OLED SEPIC PCB Layout

www.national.com 18

LM3410/LM3410Q

LM3410X SOT23-5 Design Example 1:

5 x 1206 Series LED String Application

30038581

LM3410X (1.6MHz): VIN = 2.7V to 5.5V, 5 x 3.3V LEDs, (VOUT ≊ 16.5V) ILED ≊ 50mA

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410XMF

C1, Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

D1, Catch Diode 0.4Vf Schottky 500mA, 30VRDiodes Inc MBR0530

L1 10µH 1.2A Coilcraft DO1608C-103

R1 4.02Ω, 1% Vishay CRCW08054R02F

R2 100kΩ, 1% Vishay CRCW08051003F

LED's SMD-1206, 50mA, Vf ≊ 3 .6V Lite-On LTW-150k

19 www.national.com

LM3410/LM3410Q

LM3410Y SOT23-5 Design Example 2:

5 x 1206 Series LED String Application

30038581

LM3410Y (525kHz): VIN = 2.7V to 5.5V, 5 x 3.3V LEDs, (VOUT ≊ 16.5V) ILED ≊ 50mA

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410YMF

C1, Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

D1, Catch Diode 0.4Vf Schottky 500mA, 30VRDiodes Inc MBR0530

L1 15µH 1.2A Coilcraft DO1608C-153

R1 4.02Ω, 1% Vishay CRCW08054R02F

R2 100kΩ, 1% Vishay CRCW08051003F

LED's SMD-1206, 50mA, Vf ≊ 3 .6V Lite-On LTW-150k

www.national.com 20

LM3410/LM3410Q

LM3410X LLP-6 Design Example 3:

7 LEDs x 5 LED String Backlighting Application

300385a2

LM3410X (1.6MHz): VIN = 2.7V to 5.5V, 7 x 5 x 3.3V LEDs, (VOUT ≊ 16.7V), ILED ≊ 25mA

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410XSD

C1, Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 4.7µF, 25V, X5R TDK C2012X5R1E475M

D1, Catch Diode 0.4Vf Schottky 500mA, 30VRDiodes Inc MBR0530

L1 8.2µH, 2A Coilcraft MSS6132-822ML

R1 1.15Ω, 1% Vishay CRCW08051R15F

R2 100kΩ, 1% Vishay CRCW08051003F

LED's SMD-1206, 50mA, Vf ≊ 3 .6V Lite-On LTW-150k

21 www.national.com

LM3410/LM3410Q

LM3410X LLP-6 Design Example 4:

3 x HB LED String Application

30038567

LM3410X (1.6MHz): VIN = 2.7V to 5.5V, 3 x 3.4V LEDs, (VOUT ≊ 11V) ILED ≊ 340mA

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410XSD

C1, Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

D1, Catch Diode 0.4Vf Schottky 500mA, 30VRDiodes Inc MBR0530

L1 10µH 1.2A Coilcraft DO1608C-103

R1 1.00Ω, 1% Vishay CRCW08051R00F

R2 100kΩ, 1% Vishay CRCW08051003F

R3 1.50Ω, 1% Vishay CRCW08051R50F

HB - LED's 340mA, Vf ≊ 3 .6V CREE XREWHT-L1-0000-0901

www.national.com 22

LM3410/LM3410Q

LM3410Y SOT23-5 Design Example 5:

5 x 1206 Series LED String Application with OVP

30038568

LM3410Y (525kHz): VIN = 2.7V to 5.5V, 5 x 3.3V LEDs, (VOUT ≊ 16.5V) ILED ≊ 50mA

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410YMF

C1 Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

D1, Catch Diode 0.4Vf Schottky 500mA, Diodes Inc MBR0530

D2 18V Zener diode Diodes Inc 1N4746A

L1 15µH, 0.70A TDK VLS4012T-150MR65

R1 4.02Ω, 1% Vishay CRCW08054R02F

R2 100kΩ, 1% Vishay CRCW08051003F

R3 100Ω, 1% Vishay CRCW06031000F

LED’s SMD-1206, 50mA, Vf ≊ 3 .6V Lite-On LTW-150k

23 www.national.com

LM3410/LM3410Q

LM3410X SEPIC LLP-6 Design Example 6:

HB/OLED Illumination Application

30038552

LM3410X (1.6MHz): VIN = 2.7V to 5.5V, (VOUT ≊ 3.8V) ILED ≊ 300mA

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410XSD

C1 Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106K

C2 Output Cap 10µF, 6.3V, X5R TDK C2012X5R0J106K

C3 Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

D1, Catch Diode 0.4Vf, Schottky 1A, 20VRDiodes Inc DFLS120L

L1 & L2 4.7µH 3A Coilcraft MSS6132-472

R1 665 mΩ, 1% Vishay CRCW0805R665F

R2 100kΩ, 1% Vishay CRCW08051003F

HB - LED’s 350mA, Vf ≊ 3 .6V CREE XREWHT-L1-0000-0901

www.national.com 24

LM3410/LM3410Q

LM3410X LLP-6 Design Example 7:

Boost Flash Application

30038570

LM3410X (1.6MHz): VIN = 2.7V to 5.5V, (VOUT ≊ 8V) ILED ≊ 1.0A Pulsed

Part ID Part Value Manufacturer Part Number

U1 2.8A ISW LED Driver NSC LM3410XSD

C1 Input Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 10µF,16V, X5R TDK C2012X5R1A106M

D1, Catch Diode 0.4Vf Schottky 500mA, 30VRDiodes Inc MBR0530

L1 4.7µH, 3A Coilcraft MSS6132-472

R1 200mΩ, 1% Vishay CRCW0805R200F

LED’s 500mA, Vf ≊ 3 .6V, IPULSE = 1.0A CREE XREWHT-L1-0000-0901

25 www.national.com

LM3410/LM3410Q

LM3410X SOT23-5 Design Example 8:

5 x 1206 Series LED String Application with VIN > 5.5V

30038571

LM3410X (1.6MHz): VPWR = 9V to 14V, (VOUT ≊ 16.5V) ILED ≊ 50mA

Part ID Part Value Mfg Part Number

U1 2.8A ISW LED Driver NSC LM3410XMF

C1 Input VPWR Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 Output Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

C2 Input VIN Cap 0.1µF, 6.3V, X5R TDK C1005X5R1C104K

D1, Catch Diode 0.43Vf, Schotky, 0.5A, 30VRDiodes Inc MBR0530

L1 10µH 1.2A Coilcraft DO1608C-103

R1 4.02Ω, 1% Vishay CRCW08054R02F

R2 100kΩ, 1% Vishay CRCW08051003F

R3 576Ω, 1% Vishay CRCW08055760F

D2 3.3V Zener, SOT23 Diodes Inc BZX84C3V3

LED’s SMD-1206, 50mA, Vf ≊ 3 .6V Lite-On LTW-150k

www.national.com 26

LM3410/LM3410Q

LM3410X LLP-6 Design Example 9:

Camera Flash or Strobe Circuit Application

30038572

LM3410X (1.6MHz): VIN = 2.7V to 5.5, (VOUT ≊ 7.5V), ILED ≊ 1.5A Flash

Part ID Part Value Mfg Part Number

U1 2.8A ISW LED Driver NSC LM3410XSD

C1 Input VPWR Cap 10µF, 6.3V, X5R TDK C1608X5R0J106K

C2 Output Cap 220µF, 10V, Tanatalum KEMET T491V2271010A2

C3 Cap 10µF, 16V, X5R TDK C3216X5R0J106K

D1, Catch Diode 0.43Vf, Schotky, 1.0A, 20VRDiodes Inc DFLS120L

L1 3.3µH 2.7A Coilcraft MOS6020-332

R1 1.0kΩ, 1% Vishay CRCW08051001F

R2 37.4kΩ, 1% Vishay CRCW08053742F

R3 100kΩ, 1% Vishay CRCW08051003F

R4 0.15Ω, 1% Vishay CRCW0805R150F

Q1, Q2 30V, ID = 3.9A ZETEX ZXMN3A14F

LED’s 500mA, Vf ≊ 3 .6V, IPULSE = 1.5A CREE XREWHT-L1-0000-00901

27 www.national.com

LM3410/LM3410Q

LM3410X SOT23-5 Design Example 10:

5 x 1206 Series LED String Application with VIN & VPWR Rail > 5.5V

30038573

LM3410X (1.6MHz): VPWR = 9V to 14V, VIN = 2.7V to 5.5V, (VOUT ≊ 16.5V) ILED ≊ 50mA

Part ID Part Value Mfg Part Number

U1 2.8A ISW LED Driver NSC LM3410XMF

C1 Input VPWR Cap 10µF, 6.3V, X5R TDK C2012X5R0J106M

C2 VOUT Cap 2.2µF, 25V, X5R TDK C2012X5R1E225M

C3 Input VIN Cap 0.1µF, 6.3V, X5R TDK C1005X5R1C104K

D1, Catch Diode 0.43Vf, Schotky, 0.5A, 30VRDiodes Inc MBR0530

L1 10µH 1.5A Coilcraft DO1608C-103

R1 4.02Ω, 1% Vishay CRCW08054R02F

R2 100kΩ, 1% Vishay CRCW08051003F

LED’s SMD-1206, 50mA, Vf ≊ 3 .6V Lite-On LTW-150k

www.national.com 28

LM3410/LM3410Q

LM3410X LLP-6 Design Example 11:

Boot-Strap Circuit to Extend Battery Life

30038574

LM3410X (1.6MHz): VIN = 1.9V to 5.5V, VIN > 2.3V (TYP) for Start Up, ILED ≊ 300mA

Part ID Part Value Mfg Part Number

U1 2.8A ISW LED Driver NSC LM3410XSD

C1 Input VPWR Cap 10µF, 6.3V, X5R TDK C1608X5R0J106K

C2 VOUT Cap 10µF, 6.3V, X5R TDK C1608X5R0J106K

C3 Input VIN Cap 0.1µF, 6.3V, X5R TDK C1005X5R1C104K

D1, Catch Diode 0.43Vf, Schotky, 1.0A, 20VRDiodes Inc DFLS120L

D2, D3 Dual Small Signal Schotky Diodes Inc BAT54CT

L1, L2 3.3µH 3A Coilcraft MOS6020-332

R1 665 mΩ, 1% Vishay CRCW0805R665F

R3 100kΩ, 1% Vishay CRCW08051003F

HB/OLED 3.4Vf, 350mA TT Electronics/Optek OVSPWBCR44

29 www.national.com

LM3410/LM3410Q

Physical Dimensions inches (millimeters) unless otherwise noted

6-Lead LLP Package

NS Package Number SDE06A

5-Lead SOT23-5 Package

NS Package Number MF05A

www.national.com 30

LM3410/LM3410Q

8-Lead eMSOP Package

NS Package Number MUY08A

31 www.national.com

LM3410/LM3410Q

Notes

LM3410/LM3410Q 525kHz/1.6MHz, Constant Current Boost and SEPIC LED Driver with Internal

Compensation

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